Wireless Communications Design Handbook Space Interference

WIRELESS COMMUNICATIONS DESIGN HANDBOOK Aspects of Noise, !nte~emnce, and Environmental Concerns VOLUME I" SPACE INTERF...

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WIRELESS COMMUNICATIONS DESIGN HANDBOOK Aspects of Noise, !nte~emnce, and Environmental Concerns

VOLUME I" SPACE INTERFERENCE

WIRELESS COMMUNICATIONS DESIGN HANDBOOK Aspects of Noise, !nte~emnce, and Environmental Concerns

VOLUME I" SPACE INTERFERENCE

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VOLUME !' SPACE INTERFERENCE

REINALDO PEREZ

Spacecraft Design Jet Propulsion Laboratory California Institute of Technology

ACADEMIC PRESS San Diego London Boston New York Sydney Tokyo Toronto

~ i s book is printed on acid-free paper. Copyright © 1998 by Academic Press All rights reserved. No. part of this publication may be reproduced or transmitted in any form or by any means, elec~onic or mechanical, including photocopy, recording, or any information storage and retrieval system, without permission in writing from the publisher. ACADEMIC PRESS 525 B Street, Suite 1900, San Diego, CA 92101~4495, USA http://www.apnet.com Academic Press .2.4-28 Oval Road, London ~ I

http://www~hbuk.co.ukJap/

7DX, UK

Library of Congress Catalo~g-in-Publication Data Perez, Reinaldo. "Wireless communications design handbook : aspects of noise, interference, and environmental concerns / Reinaldo Perez. p. cm. Contents: v. 1. Space interference- v. 2. Terrestrial and mobile interference- v.. 3. Interference into circuits. ISBN 0-12-550721-6 (volume I); 0-12-550723~2 (volume 2); 0~12-550722-4 (volume 3) 1~Electromagnetic interference~ 2. Wireless communication systems--Equipment and supplies. I. Title. TK7867.2.P47 1998 98-16901 621.382'24-
Contents

Acknowledgments Preface Introduction

ix

xi xiii

Chapter 1 introduction to Satellite Systems and Personal Wireless Communications t.0 1.1 1.2 1,3

Introduction Overview of Satellite Communications Launch Vehicles Launch Vehicles for the Personal Wireless Communications Revolution 1.4 Mobile Satellite Communications Overview

Chapter 2 2.,0 2.1 2.2. 2.3 2.4 2.5

Introduction Two-Body Central Force Motion. Orbital Determination KeNefian Orbits Satellite Coverage Some Te~ino!ogy

31 31 31 33 38 41 44

Attitude Control and Navigation

48

Introduction to Attitude Control. in Satellites Physical Principles of Sun Sensors Reaction Wheels:: Physical Principles Intrinsic Noise in Operational Amplifiers Star Camera: Physical Principles Noise in Amplifier Circuits Noise Sources Simple Electromagnetic Noise Coupling

48

Chapter 3 3,0 3,1 3,2 3,3 3.4 3,5 3.6 3.7

Astrodynamics of Satellites for Mobile Systems

12 15

50 55 63 64 66 69 70

vi

Contents 3,8 Common-Mode and Differential-Mode Currents 3.9 The inertial Measurement Unit

Chapter 4 4,0 4,1 4,2 4,3 4.4 4.5 4.6 4.7

Satellite Power Subsystems and Noise in Power Electronics

97 ¸

Introduction Solar Energy and Power Solar Cells and Radiation Solar A~ays The Space Environment and Radiation Damage to Solar Ceils Switching Power Supplies and ConveNers Noise in Switching Mode Power Supplies Energy Storage: Batteries

97 97 98 I02 103 105 109 t34

Chapter 5

Command and Data Handling Subsystem

5.0 5.1 5,2 5,3 5.4

Introduction Brief Satellite Command System A More Detailed View of C&DH Fundamentals of Modulation Th~ry and Coding Worst Case Analysis Guidelines for Analog and Digital Design: Examples of Use in Command and Data Handling Subsystem 5.5 Noise Issues in Satellite TeLecommunications Subsystems

Chapter 6 6,0 6.1 6,2 6.3

Noise Representations in Transponders and Multiple Access

Introduction Traveling Wave Tu~, Amplifiers in Satellite Transponders Distortions in TWTAs Multiple Access in Satellites

Chapter 7 7.0 7.I 7.2 7.3 7.4 7,5 7,6

72 74

Satdlite Antenn~

Introduction Some Fundamentals in Antenna Theory Antenna Factor and Electric Fields Antenna Interference Coupling Satellite Antenna Systems Unfurlable Antennas for Use in Mobile Communications Multibeam Frequency Reuse in Mobile Communications

141 141 141 142 148 171 193

202 202 205 208 212

222 222 222 229 230 233 242 249

Contents Chapter 8 8.0 8,1 8.2 8,3 8.4 8.5 8.6 8,7 8.8 8,9 8.t0 8,11 8.12 References Index

Space Environment and Interference Introduction Geosynchronous Environment Auroral Environment Effect of Electron Energy on Charging Spacecraft Charging Effects The Spacecraft as a Floating lhobe Charging Environments Charging Currents Differential Charging Arcing Determination of Path for Discharge Energy Circuit Upset Component Damage

Vii

251 25I 252 255 259 260 264 266 268 275 280 282 286 289 294 297

This Page Intentionally Left Blank

Acknowledgments I would like to acknowledge the cooperation arid suppo~ of Dr. Zvi Ruder, Editor of Physical Sciences for Academic Press. Dr. Ruder originally conceived the idea of having a series of three volumes to properly address the subject of noise and interference concerns in wireless communications systems. Considerable appreciation is extended to Madeline Reilly-Perez who spent many hours typing, organizing, and reviewing this book.

ix

This Page Intentionally Left Blank

Preface During the decade of the !970s and even in the early 1980s, satellite communications was basically analog (though digital was already in the drawing board), and most satellites were huge, bulky, and in geosynchronous orbit. Most of the satellites in lower orbits were either military or scientific in nature~ Things have changed within the past five to seven years. Most satellites now perform their functions using digital technology, and lower orbits such as low earth orbit (LEO) and medium earth orbit (MEO) are the main objective of most satellites being built today. These satellites are also low in weight (1000 to 1 5 ~ lb), which means that they can be launched by less expensive launch vehicles, lowering the cost not only of the satellites themselves, but also of the launch services to be used. These smaller satellites are being built for various telecommunication ente~rises embarked on bringing customers the wireless cellular and paging revolution. Most of these services will be operating by the year 2000 and are exacted to provide a huge advance for many ~ e ~ in the world lacking wireto-wire telecommunications infrastructure (about 60% of the world population!). This book, Wireless Communications Design Handbook: Aspects qf Noise, Inte~erence, and Environmental C~mcenzs, which is pan of a three-volume series, has two main objectives: (I) to provide a more detailed look at the spacecraft technology hardware itself than has been provided in other satellite telecommunications books, and (2) to redirect the emphasis in telecommunications technology to the aspects of noise and interference in wireless communications, rather than the subject well-addressed (in many other books) of digital communications technology. Many aspects of interference issues, if addressed at the hardware level, can be of great help to the designers of future satellite communications systems, especially when more compact and capable satellites are already on the drawing board, which could enhance interference problems within the spacecraft at the subsystem level~ The book is divided in eight chapters, all dealing with some aspect of interfer~ ence in a particular spacecraft subsystem. Spacecraft subsystems, of course, are reasonably well described, and they serve as a good background for the interference scenarios discussed. The description of these hardware issues really starts xi

xii

Preface

in Chapter 3. Chapter 1 is mostly dedicated to the topic of wireless personal communications in general, but there is good detail at. the hardware level about the present and future architectures being proposed in this arena, Chapter 2 serves as a preamble to Chapter 3, providing introductory material conce.rning the Newtonian mexzhanics of satellite motions. Chapter 3 addresses attitude control in a spacecraft. There is ample discussion of the hardware most. commonly used to provide the stability a satellite needs to p e r f o ~ its primary function, of providing communications services. In this chapter (as in all other subsequent chapters), we try to address some commonly obser,~ed interference and noise problems for each of the assemblies discussed. In Chapter 4 we cover the power subsystem: not only externally generated power (solar ~ a y s ) , but also switching-mode power supplies, which will become more and more common in satellite, power electronics. The interference and noise issues addressed in this chapter den mostly with switcNng power supplies. Probably the largest chapter in the book is Chapter 5, which is solely dedicated to the command and data handling subsystem. As in the discussion of attitude control, we first cover some general aspects of this space~raft subsystem, and then focus one by one on almost, all the spacecraft assemblies within this subsystem. The noise problems obse~ext in digital modulation (a theoretical approach) and the interference concerns that need to be addressed in command electronics (a hardware approach) are discussed in this chapter. The next two chapters (Chapters 6 and 7) deal mosOy with the telecommunications subsystem, with Chapter 6 addressing mostly hardware issues such as the transponder, and Chapter 7 dedicated solely to satellite antennas. It was also necessary in Chapter 7 to address several access technologies presently used in wireless personal communications and compare them from the noise point of view (this work, however, is mostly theoretical). Chapter 7 m.akes a good contribution not only to antenna theory but to special topics of antenna design for satellites; of course, interference concerns in the f o ~ of antenna coupling cases are discussed. Finally, Chapter 8 is a comprehensive chapter dedicated to space, interference and its implications for spacecraft design. This is a subject rarely covered in telecommunications books; therefore, the. need was felt to give it a good treatment in this chapter. This book is a good introduction at the hardware level to the fundamentals of spacecraft technology and interference issues of impoaance in the design of such hardware. This first volume is of a practical nature and can be taught in a classroom environment if the emphasis is on hardware technologies for spacecraft.

Introduction The information age, which began its major drive at the beginning of the 1980s with the birth of desktop computing, continues to manifest itself in many ways and presently dominates all aspects of modem technological advances. Personal wireless communication se~ices can be considered a "su..bset- technology" of the information age, but they have also gained importance and visibility over the past 10 years, especially since the beginning of the. 1990s. It is predicted that future technological advancements in the info~ation age will be unprecedented, and a similar optimistic view is held for wireless personal, communications. Over the past few years (since 1994), billions of dollars have been invested all over the world by well-known, technology-driven companies to create the necessary infrastructure for the advancement of wireless technology. As the thrust into wireless personal communications continues with more advanced and compact technologies, the risks increase of " c o ~ p t i n g " the. information provided by such communication services because of various interference scenarios. Although transmission, of information through computer networks (LAN, WAN) or through wires (cable, phone, telex:.ommunicafions) can be affected by intert%rence, many steps could be taken to minimize such problems, since the methods of transmitting the information can be technologically managed. However, in wireless communications, the medium for transmission (free space) is uncontrolleA and unpredictable. Interference and other noise problems are not only more prevalent, but much more difficult to solve. Therefore, in p~allel with the need to advance wireless corrLmunication technology, there is also a great need to decrease, as much as possible, all interference modes that could c o o p t the information provided. In this handbook series of three volumes, we cover introducto~ and advanced concepts in interference analysis and mitigation for wireless personal communications. The objective of this series is to provide fundamental knowledge to system and circuit designers about a variety of interference issues which could pose potentially detrimental and often catastrophic threats to wireless designs. The material presented in these three volumes contains a mixture of basic, interference fundamentals, but also extends to more advanced xiii

xiv

Introduction.

topics. Our goal is to be as comprehensive as possible. Therefore, many various topics are covered. A systematic approach to studying and understanding. the material presented should provide the reader with excellent technical capabilities for the design, development., and manufacture of wireless communication hardware that is highly immune to interference problems and capable of providing optimal perfomaance. The present and future technologies for wireless personal communications are being demonstrated in three essential physical arenas:, more efficient satellites, more versatile fixed ground and mobile hardware, and better and more. compact. electronics. There is a need. to understand, analyze, and. provide corrective measures for the kinds of interference and noise problems encountered in each of these three technology areas.. In this handbook series we provide comprehensive knowledge about each of three technological, subjects. The three-vo!ume series,

Wireless Communications Design Handbook: Aspects of Noise, Interference, and Environmental Concerns, includes Volume L Space Interference; Volume 11, Terrestrial and Mobile Interference; and Volume iii, Inte~erence inW Circuits. We now provide in this introduction a more detailed description of the topics to be addressed in this handbook.

Volume 1 In the next few years, starting in late 1997, and probably extending well into the next century, hundreds of smaller, cheaper (faster design cycle), and more sophisticated satellites will be put into orbit.. Minimizing interference and noise problems witNn such satellites is a high priority. In Volume 1 we address satellite system and subsystems-level design issues which are useful to those engineers and managers of aerospace companies around the world who are in the business of designing and building satellites for wireless personal communications. This material could also be useful to manufacturers of other wireless assemblies who want to understand the basic design issues for satellites within which their hardware must interface. The first volume starts with a generalized description, of launch vehicles and the reshaping of the space business in general, in this post-Cold War era. A description is provided of several satellite, systems being built presently for worldwide access to personal communication services. Iridium, Globalstar, Teledesic, and Odessey systems are described in some detail, as well as the concepts of LEO, MEO, and GEO oft)its used by such satellite systems.

Introduction

xv

Attention is then focused briefly on the subject of astrodynamics and satellite orbital mechanics, with the sole objective of providing readers with some background on the importance of satellite attitude control and the need to have a noise.-free environment for such subsystems. Volume 1 shifts to the study of each spacecraft subsystem and the analysis of interference concerns, as well as noise mitigation issues for each of the satellite subsystems. The satellite subsystems addressed in detail include attitude and control., command and data handling, power (including batteries and solar arrays), and communications. For each of these subsystems, m~or hardware assemblies are discussed in detail with respect to their basic functionalities, major electrical components, typical interference problems, interference analysis and possible solutions, and worst-case circuit analysis to mitigate design and noise concerns. Considerable attention is paid to communications subsystems: noise and interference issues are discussed for most assemblies such as transponders, amplifiers, and antennas. Noise issues are addressed for several multiple access techniques used in satellites, such as TDMA and CDMA. As for antennas, some fundamentals of antenna theory are first addressed with the objective of extending this work to antenna interference coupling. The interactions of such antennas with natural radio noise are also covered. The next subject is mutual interference phenomena affecting space-borne receivers. This also includes solar effects of VHF communi-. cations between synchronous satellite relays and e ~ h ground stations. Finally, satellite antenna systems are discussed in some detail. The final section of Volume 1 is dedicated to the effects of the space environment on satellite communications. The subject is divided into three parts. First, the space environment, which all satellites must survive, is discussed, along with its effects on uplink and downlink transmissions. Second, charging phenomena in spac~raft are discussed, as welt as how charging could affect the noise immunity of many space.craft electronics. Finally, discharging events are investi~ gated, with the noise and interference they induce, which could affect not only spacecraft electronics, but also direct transmission of satellite data.

"Volume 2 In the second volume, of this handboook series, attention is focused on system-. level noise and interference problems in ground fixed and. mobile systems, as well as personal communication devices (e.g., pagers, cellular phones, two-way radios). The work starts by looking at base station R1z communications systems and mutual antenna interference. Within this realm we address interference be-

xvi

introduction

tween satellite and e ~ h smti.on links, as well as interference between broadcasting te.~estfial stations and satellite ea.~h stations. In this approach we follow the previous work with a brief introduction to interference canceling techniques at the system level. Volume 2 devotes considerable space to base-station antenna performance. We address, in. reasonably good technical detail, the most suitable antennas for base-station design and how to analyze possible mutual interference coupling problems. The book also gives an overview of passive repeater technology for personal communication services and the use of smart antennas in such systems. A section of Volume 2 is dedicated entirely to pagers and cellular phones and interference mitigation methods. The fundamentals of pagers and cellular phone designs are studied, and the use of diversity in antenna design to minimize interference problems is reviewed. A major section of this volume starts with the coverage of propagation models for simulating interference. In. this respect we cover Rayleigh fading ~ it relates to multipath interference. Path loss, co-channel, and adjacent channel interference follows. This last material is covered in good detail, since these techniques are prevalent in the propagation models used today. The last sections of Volume 2 deal in. depth with the subject of path loss, material that needs better coverage than found in previous books. The following subjects are reviewed in detail" ionospheric effects, including ionospheric scintillation and. absorption; tropospheric clear-air effects (including refraction, fading, and ducfing); absorption, scattering, and cross-polarization caused by precipitation; and an.. overall look at propagation effects on interference.

Volume 3 In Volume 3, we focus our attention inward m address interference and noise problems within the electronics of most wireless communications devices. This is an. import.ant approach, because if we can mitigate interference problems at some of the fundamental level.s of design, we could probably take great steps toward diminishing even more complex noise problems at the subsystem and system levels. There are many subjects that could be covered in "volume 3. However, the material that has been selected for instruction is at a fundamental level and useful for wireless electronic designers committed to implementing good noise control techniques. The material covered in Volume 3 can be divided into two major subjects: noise and interference, concerns in digital electronics, including mitigation responses; and noise and interference in analog electronics,

Introduction

xvii

as well as mitigation responses. In this volume we also address computational electromagnetic methods that could be used in the analysis of interference problems. In the domain of digital electronics we devote considerable attention to power bus routing and proper grounding of components in. printed circuit boards (PCBs). A good deal of effort is spent in the proper design of power buses and grounding configurations in PCBs including proper layout of printed circuit board traces, power/ground planes, and. line impedence matches. Grounding analysis is also extended to the electronic box level, and subsystem level, with the material explained in detail. At the IC level, we concentrate, in the proper design of ASIC and FPGA to safeguard signal integrity and avoid noise problems such as ground bounce, and impedance reflections. Within the area of electronic design automation (EDA), parasitics and verification algorithms I~r ASIC design are also discussed. A great deal of effort is put into the study of mitigation techniques for interference from electromagnetic field coupling and near-field coupling, also known as crosstalk., including crosstalk among PCB card pins of connectors. The work continues with specific analysis of the interactions in high-speed digital circuits concerning signal integrity and crosstalk in the time domain. Proper design of digital grounds and the usage of proper bypass capacitance layout are also addressed. Other general topics such as power dissipation and thermal control in digital IC are also discussed. Electromagnetic interference (EMI) problems arising in connectors and vias are reviewed extensively, including novel studies of electromigration in VLSI. in the analog domain, Volume 3 also addresses many subjects. This section starts with the basics of noise calculations for operational amplifiers. Included here is a review of fundamentals of circuit design using operational amplifiers, including internal, noise sources for analysis. As an extension concerning noise issues in operational amplifiers, the material in this volume focuses on the ve~ important subject of analog-to-digital converters (ADCs). In this area considerable effort is dedicated to proper power supply decoupling using bypass capacitance. Other noise issues in high-performance ADC are also addressed, including the proper design of switching power supplies for ADC, and the shielding of cable and connectors. Finally, at the IC level, work is included for studying P&~I rectification in analog circuits and. the effects of operational amplifiers driving several types of capacitive loads. We end this volume with the study of system-level interference issues, such. as intermodulation distortion in general, transmitters and modulators, and the. subject of cross modulation. This is followed by the concept of phase-locked loops (PLL) design, development, and operation. Because of the impo~ance of

xviii

Introduction

PLL in communications electronics, considerable space is devoted to the study of noise concerns within each of the components of PLL. Finally, Volume 3 ends with an attempt to explore interference at. the level of transistors and other components.

Chapter 1

1.0

Introduction to Satellite Systems and Personal Wireless Communications

Introduction

The first satellite was launched in 1957 by the then Soviet Union. In 1958, SCORE was the first U.S. communications satellite used for voice communications. Now, 40 years later, there are close to 150 satellites in orbit, and many more. will soon be added. Satellites p e f f o ~ a variety of tunctions" fixed satellites for communications, broadcast satellites, and mobile satellites. The work they do ranges from experimental to milita~. In the past, satellites have been used for navigation, positioning, surveillance, communications (voice, data, and video), and even monitoring geological resources from the earth and atmosphere. Lately, satellites have acquired a new use, which is part of the subject of this book: wireless personal communications, including cellular communications. Most satellites behave as active relays and switching stations in space with ~ea coverage, accessibility; and bandwidth. The satellite consists of a communications payload and the spacecraft bus. The communications payload contains an antenna system and a transponder system. The spacecraft bus has various subsystems such as electrical power~ propulsion, structural, t h e ~ a l , attitude and articulation control, command, and data handling. Figure 1.1 shows a typical satellite with some communications system. Figure 1.2 shows a block diagram of the S/C bus subsystems. 1.0.i

STRUCTURE

The spacecraft structure is built to provide a stable and strong platform: for payload instruments and subsystems. Because of its rigidity, it is capable of withstanding the great mechanical stresses imposed during various phases of the mission. Structures are manufactured from very !ightwei~t materials while maintaining the necessary strength~ The most popular structural material today is an aluminum honeycomb combined with glass fibers. 1.0.2

POWER

Solar arrays are the primary power source on board most spacecraft. The solar arrays are. deployed once the spacecraft is in orbit using special gimbals and

2

i. Introduction to Satel~te Systems and Personal Wireless Communications

t9 Element S-Band Feed Gimbalted K-Band " ~ \ \ GatewayAntenna C-Band Omni.,.. \ , Conical Scan Earth Sensor Thermal Radia

19 Element L-Band Feed

S-Band Antenna .......-~q

Antenna

Solar Array,,,.~,~, Multi-layer [ "! Reaction Insulation Fine Control Sun .~ Thruster Sensor Apogee Injection Engine Coarse Sun Sensor

Figure 1.i

Typical satellite and its associated communications system.

retention and release mechanisms. ~ e solar w a y s are then oriented so that they are perpendicular to the sun's rays. During eclipse periods (i.e., sun occultations) or for lower power requirement, electrical power is provided by storage batteries which are .themselves charged from the solar arrays. Power regulation, switching, and dis~bufion are employed to distribute power to all electronic systems.

1.0.3

ATTITUDE AND ARTICULATION CONTROL

The attitude and articulation control subsystem makes sure that the spacecraft is placed in a precise orbital location and maintains the required attitude throughout its mission. The attitude control is achieved by employing a reaction-whe~l assembly to produce gyroscopic torques or reactions to thruster control. The inertial measurement unit maintains the fight attitude. The sun sensors and star trackers allow navigation in space. To achieve accuracy of a few arc seconds, advanced techniques are employed, involving gyroscope and star-sensor measurements and onboard computers.

1.0. introduction POWER BUS

Communications Payload

Command and Data Handling

Telemetry, Tracking, and Control

Power Mechanical and Articulation

Attitude and A~iculation Control

Propulsion

Thermal Control

LOCAL DATA BUS

Figure 1.2 Subsystems of a typical cormnunicafions satellite.

1.0.4

COMMAND AND DATA HANDLING

This subsystem includes telemet~ and tracking command, and control, It enables data to be sent continuously to earth, and also permits the reception from earth

4

1. Introduction to Satellite Systems and Personal Wireless Communications

of all commands necessary to control the spacecraft's functions and assignments. The subsystem also tracks the health of the spacecraft and sends commands to c ~ out various tasks such as switching the transponders in and out of service, switching between redundant units, and firing pyros. 1.0.5

THER_MAL CONTROL

Thermal control of a satellite is n~ded to achieve temperature balance and guarantee proper performance of all subsystems. Thermal stress results from high temperature effects due to the sun, as well as from low temperatures ~curring during eclipse. Heat shields, blankets, heaters., and thermal control devices (temperature sensors, heater circuits) are employ~ to protect spacecraft bus equipment. The thermal control subsystem provides a good regulation of temperature for efficient satellite performance. 1.0.6

PROPULSION CONTROL

The propulsion subsystem allows for large orbit changes, station keeping, and spacecraft attitude-control maneuvers. Usually, orbit change is achieved by the apogee, kick/boost motor. Often, orbit changes are needed because of orbit d~ay. Other propulsion thrusters use l o w - t ~ s t cold gas or liquid mono- or bipropellant systems for attitude control. If a bipropellant is selected, then the apogee boost and auxiliary propulsion system can be combined into one system. 1.0.7

MECHANISMS

Many instruments and sensors use a variety of mechanical devices, such as deployment hinges, latching mechanisrns, rotating drives, and rotational and linear pointing mechanisms. Normally, motor drives like those found in solar array drives are controlled constantly. Some other mechanisms, such as those in charge of antenna deployment and other specialty sensors, operate only once and are achieved by means of spring-loaded or pneumatically operated actuators, which are often released by firing pyros. 1.0.8

PAYLOADS

Payloads can be categorized based on the type of work or function, such as radiofrequency experiments, magnetic field measurements, plasma measurement experiment, or transponder equipment. Usually, electrical power is provided to

1.1. Overview of Satellite Communications

5

the payloads with the nominal spacecraft, bus voltage. The payloads do all the required conversion, regulation, and inversion. A standard telemetry and telecommand inmrface is provided using an interface unit which is located close to the payloads. The telemetry interface processes the signals gathered by the payloads including housekeeping, and sends them to onboard telemetry systems where the signals are directly inserted into the data being modulated and transmitted down to the ground s~tions. In the same manner, the comm.a~d information received by the receivers will reach the payloads via interface units.

1.0.9

HARNESS

The spacecraft provides electrical connections both for signal lines and for power between all subsystems and the instruments. The harness also connects the solar array power through slip rings via a power subsystem to the various spacecraft subsystems. The satellite harness includes all interconnecting cables that will interface with each. of the spacecraft subsystems, such as power telemetry, command and data handling, communications, reaction wheel control, mad solar-array articulation mechanisms. The harness also includes umbilical wiring needed for ground checkout and launch vehicle interface. The harness is usually built to withstand large bre~down voltages. Cable isolation methods axe employed to diminish electromagnetic interference. Cable isolation schemes and shielding methods are. routinely employed to minirrfize electromagnetic interference. Pyrotechnic cables are shielded and twisted and isolated from other sensitive cables within the harness. Shielding wires are widely used to minimize electromagnetic interference. Twisted-pair cables are used to minimize magnetic fields due to the current flow in wires. Redundant twisted pairs are used for all primary power and return lines.

1.1

O v e r v i e w o f Satellite C o m m u n i c a t i o n s

A typical satellite communications system is shown in Figure 1.3. in the figure, the terrestrial network can correspond to the voice, data, or video signal from a public switching telephone network (PSTN) or from a cellular network. An earth transmitting station, also known as a gateway in. cellular technology, is used for processing signals before they are transmitted to the satellite via an uplink protocol. Signals that arrive at. an earth station through a terrestrial or cellular network include voice channels, analog channels, and data from computers and other

6

1. Introduction to Satellite Systems and Personal Wireless Communications

! ,

/

!

!

Gateway

Switch

\

\

\

\

\

\

i

Gateway Termina!

Mobile Termina~ PuNic Network Switching

Fibre 1.3 Satellite communications network,

digital data sources; these signals, which are usually called baseband signals, are deterministic or random. These signals are combined into a complex baseband signal. The combination of diverse signals coming from different sources is called multiplexing. The multiplexed signal is converted to a suitable form of signal for transmission; this is done by changing the amplitude, phase, or frequency of a sinusoidal signal. This process is known as modulation and can be analog or digital, in f o ~ . Most preset modulation is done digitally. The main advantages of digital modulation are its error-correcting capability, insensitivity m noise, and nonlinearity. A digital satellite communications system is shown in Figure 1.4. The analog signals for the process of digital modulation are converted into binary data. These binary data are coded for error correction. The coded binary bits are modulated by several digital modulation, schemes using PY modulators. These signals can be subjected to further spread-spectrum modulation for possible interference rejection. The signals are then up-converted and amplified and transmitted, to the satellite via an. antenna system. These signals are sent through space, where they are subject to several kinds of losses and a~ive at the satetlite's antennas. Signals

1.1. Overview of Satellite Communications

Baseband Processing

Channel Encoding

Multiplexer

......................

7

ca, ier- ....................................... Modulation

+ Spread-spectrum Moduiat!on

nd

oo

mplification

Baseband Processing I and Down-

Demultiptexer

Source Coding

Channel Decoding

~

Spread Carrier "~~Spectrum Demodulation/ D e m o d u l a t i o n

Figure 1.4 Components of a digital satellite communications system.

from other earth stations or gateways also arrive at the same time as the satellite antenna. This simultaneous access by several earth stations to the satellite transponder for the relay is achieved by a method of multiple access. The signals are down-converted at various transponders, then amplified and transmitted back to earth stations by multiple beams. At the earth stations, the signals go through low-noise amplifiers, are down-conveNed, despread, demodulated, and decoded at channel and sourceotoobaseband signals, and then demultiplexed and sent to the appropriate destination.

1. Introduction to SateUite Systems and Personal Wireless Communications 1.2

Launch

Vehicles

Satellites are put into orbit using rockets or launch vehicles. Rockets are propelled to very high speeds into the higher atmosphere as a result of chemical combustion that exits through a nozzle. The physics of rocket motion is as simple as that of conservation of momentum. C o n t r ~ to the air-breathing engine of a jet plane, the rocket does not push against air or any other matter, which would result in an impulse function from Newton's third taw of dynamics; instead, its forward momentum is the result of gas molecules being ejected backward as a result of the internal combustion, and again by Newton's third law of motion, the launch vehicle is propelled forward. The rocket therefore is the only launch vehicle capable of traveling through vacuum. The impulse and lifting capabilities of a launch vehicle depend on its thrust, size (i.e., mass), launch trajectory, relative size of the different stages, altitude of final orbit, and accuracy of the final orbit. If we consider that thrust (which is the lift force) multiplied by the time it acts is called impulse, as given by Equation (i. 1), then limpu)se = T X At.

(I. I)

~ e acceleration of the launch vehicle is given by

( T - W) Wg

a .............................,

(1.2)

where W is the weight of launch vehicle and its satellite, and g is the constant for gravity (9.8 rrdsecZ). As the rocket fuel continuously bums during ascent, its weight gradually decreases, and with an increase in altitude, the value of g decreases. The average velocity at any instant of time during ascent is given by v(t)= ~x(t)In(M], kin/:

(1.3)

where ~x(0 is the exhaust velocity relative to the rocket, M is the mass of the rocket at launch, and m is the rocket mass at the instant when the velocity is v(t). As a multistage rocket ascends, empty stages are discarded as .their fuel runs out. The weight of the rocket decreases, and this results in an increase of velocity in ~ u a t i o n (1.3). Acceleration increases (see Equation (1.2)), and eventually this increases the impulse. It is clear from this example that the use of multiple stages represents an advantage. Because this is a relationship between altitude

1.2, Launch Vehicles

9

and velocity, it is of great, advantage for a satellite to reach optimum altitude first using a low vertical speed, then to accelerate to reach the needed horizontal velocity. This scenario can be observed in Figure 1.5. Notice from the figure and Equation (1.3)that at the beginning of ascent the mass ratio CM/m) is low and only a small rocket velocity can be achieved. As M/m increases suddenly because of release of the first stage (after all its fuel is used), the rocket speed that can be attained increases dramatically; this is in addition to the velocity previously attained:. For a third, stage, the gain is even more pronounced. However, notice from Figure 1.5 that the curve flattens out as M/m increases. Therefore, after a third stage the gain in. velocity is not that significant; this is why most launch vehicles do not go beyond a third stage.

Gravity and drag ignored

4

0

10

20

30

40

Mass Ratio Figure 1.5 Velocity vs mass ratio of an ascending rocket.

10

1. Introduction to Satellite Systems and Personal Wireless Communications

A variety of launch vehicles are used to put satellites into orbit. Figure t.6 shows schematics of the Titan IV, which is used primarily by the U.S. Air Force for launching very heavy loads (over 2 tons). Other launch vehicles of the Titan family are also shown in Figure 1.7. Figure 1.8 shows the Afiane launch vehicle family, which actively launches more than 50% of commercial satellites at present. In the United States, because of launch site and range safety considerations specific launch sites have to be considered for particul~ inclination angles. This means that satellites requiring a low inclination angle must be launched from the E~tern Test Range as shown in Figure 1.9. Such satellites are launched in a general easterly direction. Satellites requiting high-inclination or polar orbits must be launched from the Western Test Range, as shown in Figure 1.10, and in a general southerly direction.

Length Diameter Thrust Propellants

Solid Rocket Motors

Stage One

112..ff t0.0 ft. 1~6 million Ib per motor solid t . . . . . . .

85~5 ft t0.0 ft 546,000 ibs storable ~iquid . . . . . .

L~

1

326 ft 10.0 ff 104,000 lb storable liquid . . .

204 ft

I

[

Stage Two

86 ft

1~ ~

~qn|iH R n p k . ~ f R t a n ~ t ~ f

Figure 1.6 Titan IV configuration.

Centaur

29.3 ft t4°2 ft 33,000 tbs cryogenic

1.2. Launch Vehicles

11

_ 200ft 150 ft

_ 100ft

Titan I

Titan II Gemeni

Titan

Titan i¢l Agena, Titan ItlC

Titan tHD T!~antIJE Titan34

Titan 34D

Titan IV

Figure 1.7 The Titan launch vehicle family.

Ariane 4 Ariane 5 Artane I

Ariane 2

Ariane 3

4 m, diameter fairing

Larger fairitig 3rd stage stretched capacity: 10,5 tons Stretched tanks capacity: 226 tons

Strap-on boosters*

Figure 1.8 The Ariane launch vehicle family.

12

I. I n t r o d u c t i o n to Satellite S y s t e m s a n d P e r s o n a l Wireless C o m m u n i c a t i o n s .........

= = = .

:=-==

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~...... . . . . . .

......

45 ° t 4 7 '~ inclination 51 ° 62 °

AFRICA 90 °

~k

~°5° t27 °

1i5 °

135 °

inclination

Figure 1.9 The U.S. Eastern Test Range launch scenarios.

:;.:====..

:::::::::::::::::::::::

:: :....:::

.:..=..~:==::.....:::::;;;=...: ..:..........::::.: .....:..:.......:::.. :

t48 ° Retrograde

,°o,..,io.

:

305 °

.:...:

,

,

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===

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~JJ.-__~o.

300 ~

AUSTRALIA=

237°/

222°~//t 220°

1 172 °

' / /° 1 5 5 81 ° Inclination

Figure 1.10 The Western Test Range launch scenarios.

1.3

L a u n c h Vehicles for the Personal Wireless C o m m u n i c a t i o n s Revolution

The commercial business of launching commercial satellites for. wireless communications is advancing by great strides. One of the major objectives of the wireless

1.3, Launch Vehicles for the Personal Wireless Communications Revolution

13

personal communication business is to can 7 out hundreds of launches within the next. 5 to 10 years at very affordable rates (about $1000 per pound) that are much lower than the $ 1 0 , 0 ~ per pound that is the present cost of launch operations. To reach these low cost targets, extensive restructuring of launch-vehicle design, manufacture, and sen, icing needs to be. accomplished. Low-cost communication satellites for wireless personal communications demand equally low-cost launch vehicles. Such low cost could come from (I) more efficient transportation of launch-vehicle components to launch sites, (2) off-the-shelf technology for parts and electronic components, (3) decreased maintenance and servicing requirements of launch facilities, (4) streamlining the testing of assemblies and components (which could be helped greatly by the implementation of modular approaches in the design and assembly of the most complex subsystems), (5) increased reliability, and (6) streamlining the use of suppo~ personnel. In the present space race. for launching satellites, the main players are shown in Figure !. 11. The United States commands the launch systems with three different types of launchers: Delta (Boeing), Atlas, and Titan (Lockheed Martin). The French Aerospatiale agency, in conjunction with. a consortium of European aerospace corporations,

Them ~m b ~ y ~o c~pa~ ~w~ in t ~ launch b~nes~, ~ e ~ g ~ L~kheed

~ey me h e ~ y im~o~e~ i~ c ~ m m e ~ I ~ n c h ~

~--+~-. . . . "=

, ~ n e spa~e,: a

:

-=-----=::"=' "--Chinm

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i

~ead ~t France

~.~ce~ ~ g

e ~ a b ! e ~emme~a! f~u~h b u ~ n ~

Io pl~y ~ ro~e~ c~mm~c.i~

~ e Long

-: LQng March Ariane 4

!R~ss~: The b~ndaf¥ P m m ~ rocket ~t~ l.he n~we Z.e~iI rockel ~re t ~ d tO t~ U ~ f~ dOZe~S t~rmh~$ t h ~ h year 200(} ~r~ ~ o r ~ l , T~e Z,enR ~ l ~ISo be ~se~ fo~ semb~e~ I~ur~t~es

of

roton

#ap~m~ b~Q

pe~N~mcL ka~n~

!~a~that m expec~e~ ~ Ice ~n!y remoras..

~H2

Zenil

Figure 1.11 Nations in the Commercial Launch Business.

14

1. Introduction to Satellite Systems and Personal Wireless Communications

manufactures the Afiane 4 and. Afiane 5. China and Russia are also presently competing in the market with the Long March and Proton launch vehicles, respectively. The Japanese H2 rocket could also play a role later, in the next century. Rocket assembly starts from small components which are integrated into ever-. lmrger systems until the major components of the rocket are finally achieved in construction. An example of this process for assembly of a launch vehicle is outlined in Figure 1.12 for the Atl~-Centaur 122. The Centaur is an upper inertia launcher that plays a role as a third stage and directly puts the satellite into the desired orbit. Notice from the figure the sequence of mating the rocket

Fatdng encloses the saietlite

I L

~-~

satellite is positioned above inertial upper ~age

~

inertia[ upper stage booster is used for positioning the sateltitein the properorbit

I i I

~

............

~

fueltanks, o×,cl[ze,,and

I .~o,so.,~,beoo~.p,.

tO

~iquidpropulsion engines a e art

main booster

Figure 1.12 The assembly of a launch vehicle.

Launch Tower

1.4. Mobile Satellite Communications Overview

15

engines to the launch vehicle structures and the parallel assembly of the Centaur stage. Final assembly is performed at the launch sire, including the satellite itself. The launch sequence is described in Figure 1.13. After liftoff and almost 3 minutes into the flight, the main thruster engines cut off and the sustained engine keeps burning until the nose cone is jettisoned (.about 100 miles altitude).. Later, the sustainer engines cut off and the Centaur engine ignites. R e Centaur engine has a lower "kick," and it will rake about 5 to 10 minutes to put the satellite. into the desired "parking" orbit. From there., the satellite's own booster propels the spacecraft to the desired orbit; once there, solar arrays and antennas deploy. Finally, the satellite is "turned on." A check of all its functionalities commences, and finally the satellite is commanded to start perfo~xning its duties.

1.4

Mobile Satellite Communications Overview

Those who are newcomers to the wireless communications revolution may have the mistaken idea that the essential, role of satellite communications in this evolving industry has recently been realized. In reality, satellite communications have been around since the early 1960s, when the technology to launch and build such satellites was first developed, at the height of the Cold War. It was during these days that considerable resources were put into the development of a variety of launch vehicles, originally to carry nuclear warheads thousands of miles away, but also to put into orbit a variety of military, commercial, and spy satellites. The stam-of-the~art technology in those days was in the development of lighter, more compact and power-efficient electronic assemblies for such satellites. Highly integrated circuits were developed to perform a variety of functions which were only needed in those days for advanced applications in satellites and other kinds of spacecraft (e.g., Apollo and inte~lanetary spacecraft). More than 25 years later, satellite technology is still advancing in great strides, except that this time the advances are. driven by the expanded wireless communications market, !n the early days of communication, and especially for commercial satellites, most satellites were placed in geosynchronous orbit (GEo) around the earth's equators (in contrast m spy satellites, which mostly are placed in polar orbits). At higher latitudes, GEO satellites are observed at low elevations because of zero-degree orbit inclination. Subscribers may experience signal blockage by terrain, vegetation, or buildings. The launch vehicle vel~ity needed to place such a satellite in orbit is equivalent to that of earth rotation. At 30,000 km altitude, the satellite velocity matches that of the earth., which allows the spacecraft

16

1. Introduction to Satellite Systems and Personal Wireless Communications

Launch vehicle releases inertial upper s~ge with satellite. "

Laur¢~ veht~e takes _ sateJliteto a parking ~

Inertial upper stage puts satellite in prescribed orbR

~ '" , / / " ,~,,r ,~.~..= ~ -"--~,~

~

~

,,~

Fairing

Satelliteis turnedon V and d~oys solar a~ays AHsystems and in~mments ~ ~ J31Tm are checke¢ out "~'~~-~.~J.~ inertial Upper S t a ~ O~dizer

..............................................................

~.

~D

Pressurize

o .......

Take off

Launch Tower

;~

Fuel .................................................... . . . . . . . . .

d _

_

_

_

_

Figure 1.13 (a) The placement of a satellite into orbit.

to be fixed over a predetermined spot. Large launch vehicles (three stages needed) were used for such endeavors at a cost of $50 million to $1 ~9 million per launch. To offset the cost of launch vehicles, communications satellites for TV and telecommunications were built relatively cheaply. They basically perform their functions with a series of uplink amplifiers and a similar series of downlinks that function as broadcast repeaters.

1.4. Mobile Satellite Communications Overview

navi~aqo~a{ #am~s or #sf~ec~.o~anlensa:s ~o:r

17

a) Oom~r~d ar~ F£a/~aHandling c} P~we~ 8~bsys~em

depk~y~ds42}a~a~m/s pfevide ¢ha~#i~9~te{~t ~#

~:hnss~e~s,lo~ a~ti~U@ and

g~m~.aisfor s~lax a~ays and r~.ea~r an~aas~ Ga~.e:waybase sta~n where i.~ansmi~9 and recev{:ag s~4~als ~o#~orr~~@!ites ale p~xc,~sed

~sit~n~3

Mobile g,~nd ee~ula~ system

Figure 1.13 (b)

1.4.1

Orbiting satellite positioned and operating..

ADVANTAGES AND DISADVANTAGES OF GEO ORBITS

Because GEO satellites are fixed at a point in space above the earth, orbital "real estate." is at a premium, especially in Europe, Japan, and the United States. Closely parked satellites using the same frequency can experience crosstalk and coupling. International regulations require a 2 ° spacing for satellites that use the same frequency, and 90 ° separation for satellites using direct broadcast. GEO launches are costly, since they l~squire three-stage rockets which propel the satellites from Iow to high orbit. Finally, the 4 4 , 0 ~ - m i l e round trip of the satellite's

18

1. introduction to Satellite Systems and Persor~l Wireless Communications

signal means a time delay of 250 msec between sender and receiver: This number goes considerably higher if more than one link is needed. The main advantages of G E t satellites are that the earth station is a fixed target, transmission power requirements do not have to be high, and antennas do not have to be highly directional.

1.4.2

ADVANTAGES AND DISADVANTAGES OF LEO ORBITS

Low earth orbits (LEts) can also be effectively used for satellite communications. LEO orbits range from 250 to 10e.g)miles and signal time delays are of the order of only 5 to I0 msec. Link ~ power budgets can be as low as 1 ~ roW, with average power needed on the order of 0.5 W. Because of the lower orbit, launch costs are also diminished, since smaller or less powerful launchers will be required. However, LEO orbits also present some problems. First, a satellite in LEO orbit is a moving target for a ground station. To keep transmitted power requirements low, both the satellite and the earth station antennas must be highly directional. Furthe~ore, for LEO satellites, orbital periods are of the order of 90 minutes, which means that satellites will only be visible from 5 to 20 minutes per orbit (depending on altitude). In wireless communications satellites, "store-and-dump" techniques such as those used in weather and surveillance satellites cannot be used. Store-and-dump systems use well-designed orbits which are synchronized with earth rotations so that ground tracks are swept, eventually coveting large surfaces. The data is downlinked when the satellite passes over a ground station. In proposed continuous communications satellites for wireless technology, not only must a good set of ground stations be provided, but also such satellite networks must be capable of handling off-subscriber connections between satellites eve~ few minutes as they appear and disappear in the horizon. In Figure 1.14, we see three different kinds of proposed LEO orbits for wireless communications. In Figure 1.14a the Iridium proposal consists of six orbital planes, each plane containing 11 sa~llites. In Figure 1.14b, the Globals~r system contains inclined orbits and will need only 48 spacecraft. Finally, the Odyssey system in Figure 1.14c uses a medium earth orbit ( 1 0 , ~ kin), providing global coverage with i2 satellites and fewer earth stations ( ~ o w n as gateways). Advances in LEO satellites have become possible due to some significant improvements in electronic design, components, and power consumption. Such advances include more advanced microwave receivers and more sensitive lownoise amplifiers and receivers which can work with lower field strengths, reducing the need for higher power transmission. GaAs FETs and solid-state power

1.4. Mobile Satellite Communications Overview

19

Figure 1,14 Proposed LEO orbits in wireless communications. (From M.A, Sturza, "Satellite Systems," Microwaves & RE Nov, t995. Used with permission from Microwaves & RE)

20

1, Introduction to Satellite Systems and Personal Wireless Communications

amplifier improvements can further reduce the size of spacecraft. Systems on a chip have further reduced the size of electronic boards, the size of assemblies, and overall size of spacecraft. High-power microprocessors have added great capabilities to spacecraft attitude, articulation control and command data handling, also allowing smaller spacecraft to be built. 1,4,3

THE I R I D I U M S A T E L L I T E S Y S T E M

Motorola's Iridium system (in cooperation with Hughes aircraft) is composed of 66 LEO spacecraft, The plan calls for 6 groups of 11 spacecraft. Each group will be in its own polar orbit at 82.5 ° inclination and an altitude of 423 miles, as shown in Figure 1.15, Each plane will have 12 spacecraft in. standby mode as spares. The GSM digital cellular technology will be used. The satellite will communicate with the ground using QPSK modulation and a time-division multipie access (TDMA) protocol supporting up to 4 connections per channel. Iridium craft will talk over the microwave L band (1.6 GHz) employing phase a~ay antennas. These highly directional antennas will cover a swath of 2000 miles . ~, ?.,.:i.i: @,i~:~. 7,',. '. i ?:,...~

!:~/::i!!if+~

Figure 1.15 ~ e Iridium satellite coverage pattern. (From M.A. Sturza, "Satellite Systems," Microwaves& RE Nov. 1995. Used with permission from Microwaves & RE)

1.4. Mobile Satdlite Communications Overview

21

divided into 48 overlapping cells as shown in Figure 1.15. Any cell will support 256 simultaneous conversations. Each satellite will have a total capacity of 1100 calls. If a caller is in the middle of a conversation, an onboard digital switch transfers the caller to the next adjacent cell within the same antenna coverage. as the satellite moves out of range. If the call is still on, a contact is established with the next spacecraft approaching the horizon (it will know which satellite this is). Before the hand-off to the next samllim, the user's handset tests the new satellite connection for the appropriate signal strength and bit error rate. Satellites bundle all the traffic and downlink it to an earth station or "gateway" via a 28GHz link. If such a down!ink cannot be established, a side-looking antenna passes the traffic to adjacent satellites via a trunk, channel; other satellites can be used until the signal can be downlinked. Once the signals are received and processed, the gateway will route the signals via the. wire PSTN using standard telephone switches. Iridium handsets will require about 3 ~ mW of RF power to talk to a LEO satellite, h e y will be configured as dual-mode transceivers, enabling them to talk to the local cellular structure also (two sets of electronics are required in the front end).

1.4...4 THE GLOBALSTAR S A T E L L I T E S Y S T E M A separate satellite system sponsored by space systems Loral and Qualcomm will introduce a fleet of 48 satellites a~anged into 8 orbital, planes of 6 spacecraft each. The satellites, though in LEO orbits, will be about 1400 km above Iridium satellites and will have an inclination of 52 ° as shown in Figure 1.I6. Because of this, inclination coverage will be very low noah and south of the 77 ° parallel. By sacrificing coverage in the most northern and southern territories of the each, the G.lobals~ system guarantees at least two-satellite coverage, reducing the possibility of signal interruption m the line-of-sight microwave link. The access protocol to be used is code-division multiple access (CDMA), which allows multiple users to use the same spectrum band. The 16 fixed beams in the spacecraft antenna pattern can. handle about 1000 calls at a time. Users will talk to the Globalstar satellite system using CDMA-based handsets in the L band (1610 to 1623 MHz) and the S band. Unlike some other systems, such as the Mdium system, there is no switching; Globalstar acts only as a bent pipe or channel information in and out of a region's existing PSTN infrastructure as shown in Ngure 1.17. Even calls between two nearby satellite phones wilt have to go through the satellite's C band (.6484 to 6675 MHz) downlink, and then

22

1. Introduction to Satellite Systems and Personal. "Wireless Communications

inclined Orbits Accommodating 48 Spacecrlt

EaCh

Figure 1.16 The Globalstar o~ital diagram.

through the local PSTN's terrestrial gateway, and back m the satellite for rebroadcasting. Because of this approach, Globalstar saves resources in the extra electronics required for switching and satellite-to-satellite links. 1.4.5

THE TELEDESIC SATELLITE SYSTEM

In an effort to develop seamless near-wirelike connections for voice, video, and data anywhere and at any time, the Teledesic system has been conceived by

1.4. Mobile Satelfite Communications Overview

t:

23

.g~~

g GOCC = ground opera[ion~ cor~tro! cen|er; PLMN = punic ~ d mobile network PSTN = punic switched telepSone netwerk;

P ~ = postal, telegraph, e ~ te~ephor~e; SOCC = sa~e,lltte eperatio~ c~ntro~ center.

Figure 1,1.7 R e Globalstar system architecture. (From M.A. Stur~, "Satellite Systems," Microwaves & RE Nov. 1995. Used with pe~ission from Microwaves& RE)

Microsoft and other partners. This constellation consists of 840 interlinked LEOSATS operating in flocks of 40 spacecraft each, in 21 different polar orbits (from 695 to 705 km above earth). It will use the Ka band (30 GHz uplink and 20 GHz downlink) to provide the digital connection needed of 16 kbit/sec (minimum) to 2.048 Mbiffse~. The satellites being designed will support 1.00,~0 16-kbit channels at a given time, and each basic channel will have an associated 2 kbit/ sec D-channel used for control and signaling. Following an ISDN model, basic channels can be added in 16-bit segments to support higher data rates. Because this will be a tightly packed constellation, it will provide overlapping coverage m most of the United States, Europe, Asia, and Africa. A satellite coverage area will ~ a circle of 700 km across with a 30%: overlap from other neighbonng satellites; hence, the United States will be covered by 18 satellites at any given time. The satellite system uses: an earth-fixed cell approach in. order to accommodam the earth's relative motion. In this approach, a spacecraft covers its service area using a steering antenna, allowing the higher scanning beams to track fixedcell (52 km 2) locations on the earth. The beam stays on a fixed location on the ground until it is time to hand coverage off to another satellite. Small clusters of cells can be supported by a single beam, as it sweeps through each of them in a fixed sequence. Downlink power will be adjusted from 1 to 49 W depending on line of sight and atmospheric conditions.

24

1. Introduction to Satellite Systems and Personal Wireless Cowanumcations

The network uses fast packet switching, which is based on asynchronous transfer mode (ATM) technology as shown in Figure 1.18. Each satellite is a node in the packet switch, network, and contains intersatellite communication links with eight neighboring satellites, h i s interconne~tion arrangement forms a nonhierarchical geodesic network which is tolerant to fault and local congestion. All communications within a network are treated as streams of short fixed-length packets. Each packet contains a header, which includes an. address and sequence information; an e~or-control section used to verify the integrity of the header; and a payload section that carries the digitally encoded video, voice, and data. Conversion to and from the. packet format takes place in the terminals. The earthfixed cell design minimizes hand-off in this network. The earth surface is mapped. into a grid of 20,000 supercells arranged in trac~ parallel to the equator. Each supercell measures 160 × 160 km and is divided into nine individual cells. There are approximately 250 supercells in the band at the equator, and this number {r~tc#s~4}ileUa~s

/

....

Ope~ionsSu~por~ DataBases "~

Gateway;/Swath

~

G!gaLi.nkTe~m.in~l

Pub,tNNetwork S~tches

Figure 1.18 Packetswitching technology in Teledesic LEO satellite system. (From M,A. Sturza, "Satellite Systems," Microwaves & RF, Nov, 1995. Modified and used with permission from Microwaves & RE)

1.4. Mobile Satelfite Communications Overview

25

decreases with increasing latitude, A satellite footprint includes a maximum of 64 supercells or 576 individual cells. The number of cells for which a satellite is responsible varies as a result of orbital position and distance from other nearby satellites. The satellite closest to the center of a supercell has coverage responsibility. As a satellite passes over the earth, it sends its antenna beams to the fixed cell locations within its footprint as shown in Figure 1.19, The Teledesic system combines space-division multiple access, TDMA, and frequency-division multiple access as shown in. Figure 1.20. At any time, each supercell is served by one of ~ transmitted and one of 64 received beams from a satellite. Operating with a 23,111-msec scan cycle, a scanning beam scans the inner cells within the supercell and supports a 1440kbidsec channel, FDMA is used for the uplink, ~ d TDMA is u s ~ for the downlink. The system's communications links provide data, video, and voice as 512-bit packets. The uplinks exercise control over the RF transmitters so that only a

js,

\\ \\ / \

\ \

/

\ / \

X

'-1~9

I . . . . .

/

I

/

1 i

/

4

I-- -2--~-31 SUPERCELL 8-

/

i I I I

Frequency management can be reduced by using beam steering.

/

' 14 I 5

Cell Figure 1,19 Frequency management and satellite beam steering.

26

1. Introduction to Satellite Systems and Personal Wireless Communications Time 2

Time 3

Time I

&

CELL SUPERCELL

Fig~are 1.20 Cell scanning process in the Teledesic system. (From M.A. Sturza, "Satellite Systems," Microwaves & RE Nov. 1995. Used with pe~ission from Microwaves & RE) reasonable amount of power is used for the desired communications. The average transmitter power ranges from 0.01 to 4.7 W, depending on antenna diameter, channel rate, and atmospheric conditions. In its service area, each satellite supports a combination of up to 100,000 simultaneous basic channels. Operating in the 60-GHz band, intersatellite links connect each satellite with its eight neighboring satellites. The deployed satellite is 12 m in diameter, and the solar ~a2¢ measures 12 m on each side. There are three large panels containing phase m a y antennas. The octagonal base plate suppogs eight pairs of ISL antennas. The h e ~ of the teleco~unications subsystem is the fast packet switch, as shown in Ngure !.21. The fast packet switch routes packets to and from the scanning beam, the gigalink satellite link, and the intersatellite links transmitters and receivers. The fast packet switch features a throughput of more than 5 Gbit/sec. The frequency reference subsystem gives a stable frequency and time reference. The computer subsystem provides control information to the system. The scanning beam subsystem consists of 64 t r a n s i t channels and 64

1.4. Mobile Satellite Communi~tior~ Overview

27

Fast Packet Switch

Moduta.tor

Demodulator Decoder

._•

computer subsystem

Figure 1.21 Fast packet switching of a Teledesic system. (From M.A. Sturza, "Satellite Systems," Microwaves & RE Nov. 1995. Modified and used with permission from Microwaves & RE) independent receive channels. Each transmit channel receives digital data. packets from the fast. packet switch. The packets are encoded and modulated to form an intermediate frequency. The IF is up-converted and applied to an active element phased-array antenna that inco~orates a GaAs MMIC power amplifier. ~ e antenna converts the RF signal to an. electromagnetic field properly polarized for the earth fixed ceil. The RF signal is down-converted to an IF signal, de.modulated, and decoded. 1.4.6

THE ODYS S E Y S A T E L L IT E S Y S T E M

The Odyssey constellation of I2 satellites will orbit in three 55 ° inclined planes of 10,354 kin. Two satellites will be visible to subscribers anywhere on earth and at any time. This dual visibility provides high line-of-sight elevation angles, which minimizes obstruction by large earth objects such. as mountains, buildings, and trees. Each satellite generates a mulfibeam antenna pattern for a set of adjacent cells. The communications protocol used for both uplink and downlink is CDMA. This method allows the sharing of bands with other systems and applications. ~ e Odyssey will have satellites in MEO orbit and will provide a link between mobile subscribers and the public switched telephone network. The user circuits

28

1. Introduction to Satellite Systems and Personal Wireless Communications

with a particular satellite will enter the PSTN at a ground station located within the region s.e~ed by the satellite. If a region is shared by two or more satellites, multiple ground stations will be required. The satellite antennas are designed to provide coverage to only a portion of the total that are visible by the satellite. The satellite attitude is controlled so that the antenna remains pointed in. the designed direction. Therefore, antenna steering, as in the Teledesic constellation, is essential. Uplink transmissions from subscribers to the MEO satellite are conducted at L-band (1610 to 1626.5 MHz), while downlink transmissions are at S-band (2483.5 to 2 5 ~ MHz) as shown in Figure 1.22. Each cell of the satellite antenna pattern will be assigned a pair of frequency subbands covering about one-third of the allocations in the uplink and downlink directions. Transmissions between the ground station and the satellite take place, at Ka bands. Distinct subbands are reserved for the transmission to and from each cell. In the return direction, the composite signals received from the different cells are frequency-division multiplexed before the translation from the L band to the Ka band. in the forward direction, the satellite demultiplexes the FDM uplink transmission into its subband signals following translation from the Ka to the S band. The composite subband signals are routed to the v~o.us downlink antenna feeds. The required Ka-band bandwidth in either the forward or backward directions is the product of the subband bandwidth and the number of cells in the spacecraft antenna pattern. Call routing priority will be provided primarily by terrestrial cellular services. When a call is placed, the handset looks for a cellular frequency and attempts to place a call in the local cellular network. If cellular service is not available, the call will be refe~ed to the beam which provides the strongest signal to the earth station, and will be instructed to use the frequency and a particular spreadspectrum code appropriate to that beam. Most of the time the user will remain within the same ceil. ( 8 ~ ~ in diameter), but in long calls the user can. be reassigned to a different beam. If the call is long enough, rerouting to a different satellite will be arranged by the base station.. As previously stated, Odyssey is a spread-spectrum signaling method, and this allows the. frequency spectrum to be shared by multiple service operations. Spread spectrum also reduces the data rates and power for signal transmission. The Odyssey handset will be slightly larger than a regular cellular phone because it will operate at both cellular and satellim frequencies; the antenna will be quadfifilar helix and will transmit an average of 0.5 W. Because of their MEO location, Odyssey satellites will have a greater antenna gain to compensate for the greater path loss, rain, vegetation, etc. The high. elevation angle of 30 ° also contributes to antenna efficiency. The Odyssey design inco~orates t9-channeI architecture for both f o ~ a r d and remm links. In

1.4. Mobile Satellite Communications Overview L-Band Receiver

L-Band to tF Down Conye..~r . . . . ...... , L i m , e r . . ~ 7 ~ i l } .

Antennas

•"

----

~

U/C~~,

~NLE.!/I.. D_!c

....

IF tO Ka

Fi~ter & Combiner

.

29

- ..........

I

Fitter & Combiner

IF Io Ka

i

~~[IIID:'C._ILi~i. i.ii~.~i..}-~ ~ witch t8 for 25 R~undant

~-~

I~ Ka-band Antenna TxfRx

~

..... ! Switch 25 to 19, Redund

_.~B,~

AntennaTx/Rx

Ka-3"~ I

Bano~ws;;A ~~ Ka*Band LNA

-

...

.. BPF

Ka~Band LNA

411-{..,;~i.~-Am~ss~ ~... A~

...................

,ql--ID~i Amp/~PAjmqI-~ i .......................... ...........

!

•

lZ i

• •

i

/

S~Band MUX :

........................

~ c ; .........~_J, ..

B~tch 25 for 19 Redun ta.nt S~Band

S-BandSSPA

L. . . . . . . . . . !

Switch 19 for 25 Redundant

! Tc F - K - d

t

~

~

"~I

TX,

3

S-Band MUX

..

A

"r

Ka* 3and 7 way !

L014

Antenna

Figure 1.22• The Odyssey satellite telecommunication system.

30

1. Introduction to Satellite Systems and Personal Wireless Communications

the return link, each of the 19 received beams are sent to a low-noise amplifier (LNA), unconverted to Ka band, amplified by a solid-state power amplifier (SSPA) or traveling wave tube (TWT), and then directed to the Ka-band base station antenna. The forward link wilt be the complement, of the return link.

Chapter 2

2.0

Astrodynamics of Satellites for M o b i l e S y s t e m s

Introduction

One of the most essential subsystems of any satellite or spacecraft is the attitude and articulation control subsystem (AACS), sometimes called the guidance subsystem. The AACS is responsible for proper spacecraft navigation; its sensors are tuned to scan the heavens, which serve as a reference map for the spacecraft trajectory. The AACS is also responsible for providing inertial measurements which will be used to maintain the satellite in a proper balance so that it can perforrn its basic function of serving as a communication platform. Each AACS sensor is a sophisticated control assembly with the sole mission of guaranteeing the proper satellite attitude. In this chapter we will explore briefly some of the noise and interference problems which are common to AACS assemblies and how we can diminish the risks associated with such interference problems. However, we will start this chapter with some fundamental concepts of space, mechanics which will help us later understand the inner working of AACS sensors.

2.1

Two-Body Central Force Motion

Consider the motion of a body of mass m under a central force as described in Figure 2~1. The vectors, r and 0 are relat~ by the equations r=xcos

0+ysin

0

0 = - x sin 0 + y cos 0. It follows from Figure 2.1 that dr ...... 0, dO

dO --= dO

-r~

The motion of a mass in polar coordinates is described by

dr dt

v~t) . . . .

dr dt

31

r+ r

dr dO dO d t

(2.1) (2.2)

32

2. Astrodynamics of SateRites for Mobile Systems

y .4L 0

,i"

/

r

m

,i, I

lb....

X

Figure 2.1

Motion of a mass m under a central force.

The acceleration vector is 2

(dZr (dO)) ( dzO dr dO) a(t). = \,dt 2 - r -d7 r+ rd~_ + 27~ O,

(2.3)

a(t) = rat + Oao.

(2.4)

where

The t e ~ r(dO/dt) 2 = v~/r is called the centripetal acceleration arising from the motion in the 0 direction. If the expression is d 2r/dt2 = dr~dr = 0, the path is a circle, and ar = -v2o/r. The term 2(dr/dt)(dO/dt) is often called the coriolis acceleration. Because the presence of acceleration implies the presence of a force F(t) =

m(rar + Oao).

(.2.5)

From ~ e conse~ation of angular momentum, and assuming that the motion is on one plane,

d 2r m~-

[ d,\O2 mr,--d-f)= F(r)

d 20 & dO _ O. m r - ~ + 2m & dt -

(2.6) (2.7)

2.2. Orbital Determination

33

Multiplying Equation (2.7) by r and integrating, we obtain dO m r 2 - 7 = L = a constant, dt

(:2.8)

where L is the angular momentum. Integrating Equations (2.6) and (2.7) and using Equation (2.8), we obtain 2

+ V(r) = E

m F 2 -St

1

dr

2 + V(r) = E,

where E is the energy constant and V(r) = - f F ( r ) integrating, we have

ji ';.

dr E-

(2.9)

(2.10)

dr. Solving for d r / d t and

(2.11.)

V(r) - 2.mr,~

Integrating Equation (2.8), we also obtain

O=<+f dt t

0

2.2

mr 2

(2.12)

Orbi~l Determination

We have obtained the solution of Equations (2.6) and (12.7) in terms of the four constants L, E, i~, 0o. From Equation (2.8) and substituting into Equation (2.6), d ~-,r m dt 2

L2 mr 3

= F(r), ,

or, after some manipulation, d2r L2 m-~ ~ F(r) 4.............. m r 3.

(2.13).

34

2. A s t r o d y n a m i c s o f S a t e l l i t e s for M o b i l e S y s t e m s

This equation has the f o ~ of an equation of motion in one dimension for a mass subject to the actual force F(r) plus a centrifugal force LZ/mr 3. The effective potential energy corresponding to the force F(r) is given by Veff~tive = -

sF(r) dr -

dr = V(r) + 2mr2.

(2. I4)

The integral, in Equation (2:.11) is difficult to eva!uam and the resulting equation difficult to solve for r(t). It is sometimes easier to find the path of the mass in space than to find its motion as a function of time. The resulting equation becomes simpler if we m ~ e the substitution U =

1 --, r

1 u

or

r = -

Then, using Equation (2.8), we obt~n

dr dt

1 du dO = _r2dO du u 2 dO dt dt dO L du redO

and

d2r dr. 2

L d2u dO = dO 2 &

L2u 2 d2u m 2 dO2"

m

(2.15)

Substituting for r mad d2r/dt 2 in Equation (2.13) and multiplying by -m/(L2u2), we have a differential equation for the path or orbit in terms of u(t)" m

d--~ = - u - ~ F

.

(2.16)

In the case of a mass moving under the action of a central force inversely proportional to the square of the distance from the center, k F(r) = 7 r.

(2.17):

For gravitational forces, for example, k = - G m l m 2, where G = 6.67 × 10 -s dyn/g cm z. Combining Equations (2.16) and (2.17), we have

d2u d0 2 + u =

~

~

mk ~.

(2.18)

2.2. Orbital Determination

35

The solution of Equation (2.18) is given by 1

mk

r

~

u = - =

+ A(cos( 0 - 0o)),

(2.19)

where A, ~) are arbitrary constants. Equation (2.19) is the equation of a conic section (ellipse, parabola, hyperbola) with focus r = 0. The constant 0o determdnes the orientation of the. orbit in the plane. The constant A determines the turning points of the r motion, which are given by 2

_~_+

r2 =

Lz

+2mE]

-~-

+/--~-j

ti2

(2.20a)

,

(2.20b)

which are solutions for a given E of Equation (2.14) for F(r) given by Equation (2.17). By comparing, we can conclude that A in Equation (2.19) is given by

m2k a =

--~+

2mE]1/2 L2 ]

.

(2.21)

An ellipse is defined as the curve traced by a mass moving so that the sum of its distances from two fixed points p and p' is constant. The points p, p' are called the foci of the ellipse. Using the notation of Figure 2.2, we have r' + r = 2a

(2.22)

ae

P

I bl

X a

Figure 2.2 Conic section representation of an ellipse.

36

2. A s t r o d y n a m i c s of Satellites for M o b i l e S y s t e m s

where a is half the major axis of the ellipse. In terms of polar coordinates and the cosine law, r ' e = r e + 4aCe 2 + 4 r a e cos 0,

(2.23)

where as is the distance from the center of the ellipse to the foci; e is called the eccentricity of the ellipse. If e - 0, the ellipse becomes a circle.. As e - ~ I, the ellipse degenerates into a parabola. Substituting r' from Equation (2.22) into Equation (2.23), we find a ( 1 - e 2) r .............................. t + e cos 0

(2.24)

A hyperbola is defined as the curve traced by a m ~ s moving so that the difference of its distance from two fixed foci p , p ' is constant as shown in Figure 2.3. A hyperbola has two branches defined by r' -

r = 2a

r'-r=-2a

(+ branch) (-branch).

The equation of the hyperbola becomes, in p o l ~ coordinates, a(e e - 1) r = + 1 + e cos 0"

(2.25)

D a)

~

/t..

/ o ...........1.. o ...............

I ..............................2 ~ ! /

i a~

b)

~ ..................... x

. . . . . . . . . . . . . . . . . . . . .

+branch

X- branch

Figure 2.3

, , , , , ,i...._. .

........ X

i-.,q ...../ " ..................,.. . . . . . . . . .

____j

Geometry of hyperbola and parabola.

2.2. Orbital Determination

37

A parabola is the curve traced by a mass moving so that its distance from a fixed line D (the directfix) equals its distance from fixed focus p. From Figure 2.3, we have a

i -

(2.26)

1 + cos 0

We can write the equation for all tDee conic sections in the form i - = B + A cos 0,

(2.27)

r

where A is positive, and B and A are given as follows: B > A, ellipse: 1

B

g

a(i - e2) '

A

a(1 - e 2)

B = A, parabola: 1

B--

a

A=-.

1 a

0 < B < A, hyperbola ( + branch): B -A < B<

1 a(e 2 -

I)'

A

a(/¢ 2 -

1)"

0, hypefbola ( - branch): B=-

a ( e 2 -- t ) '

A=

a(g 2 -- 1)"

If we allow an a r b i t r ~ orientation of the curve with respect to the x-axis, then ~ u a t i o n (2.27) becomes 1 -= r

B + A cos ( 0 -

0o).

(2.28)

Notice the simSlarity of Equation (2.28) to Equation (2.19). For all cases, A e - t-=r,

(2.29)

38

2. Astrodynamics of Satellites for Mobile Systems

and for all ellipses or hyperbolas, a

(:2.30)

=

From Equations (2.19) and (2.28) through (2.30), we finally obtain B=

-mk

L2

A =

B2 +

e =

1 +

(2.31)

L /

(2.32)

2 EL2..'~ 1/2 mk 2 ] .

(2.33)

For an attractive force (k < 0), the orbit is an ellipse, para~la, or hyperbola, depending on whether E < +, E = 0, or E > 0; if a hyperbola, it is a + branch. If we have a repulsive force (k > 0), then E > 0 and the orbit can be only a - branch of the hyperbola. For elliptic and hyperbolic orbit, a

=

(2.34)

The maximum and minimum radio of an ellipse ease rl,2 = a(1 +_. e),

(2.35)

and the minimum radius for a hyperbola is r~ = a(e + I).

(2.36)

Comparing Equations (2.35) and (2.36) with Equation (2.20), we can deduce that if we know that the orbit is an ellipse or hyperbola, we can find the size and shape of orbit from Equation (2.20) with the help of Equations (2.35) and (2.36),

2.3

Keple~an Orbi~

Kepler's laws were formulated ~ f o r e Newton's postulates concerning motion. These laws were determined from observations of the sun and planets. Kepler was able to conclude that the motion of planets is in the form of conic sections,

2,3. Keplerian Orbits

39

as we were abte to prove from. Newton's laws in the previous section. Kepler's laws are as follows: I. The orbit of each planet is in the shape, of an ellipse with the sun at one of the foci. 2. The radius vector from the sun to a planet sweeps out equal areas in equal times. 3. R e square of the period of a planet's orbit is propo~io.nal to the cube of its semimajor axis. ~ e s e laws are the results of the gravitational central force inverse square, taw of Figure 2.4. Since in Equation (2. i0) the second tenu is a constant based on the conservation of angular momentum, we can rewrite such an equation as l

~,

k

E = ~ mv~(r) "

r

=

k

.

2a

(2.37)

From Equation (2.37), we have

v 2=k2

m(r -

~)

(2.38)

'

V(r)

ro I

• . K
. ,

-

r .

"

1/2(K2mL 2 ) Figure 2,4

Effective potential for central inverse square law.

40

2. Astrodynawdcs of Satellit~ for Mobile Systems

which is known as the vis-viva equation and is the equation for the velocity of a satellite of mass ms. For a satellite above the earth, the equation can be simplified considerably knowing that k = G mem s and m becomes m s, where m e is the mass of the earth and m s is the mass of the satellite. Assuming that r = r e + h, where r e is the radius of the earth (6378 km) and h is the instantaneous altitude, Equation (2.38) b ~ o m e s v2 = 398,60016382 + h

2.3.1

~].

(2,39)

CIRCULAR ORBIT

For circular orbits, e - 0 in Equation (2.24):, which means that r = circular). Therefore, Equation (2.38) becomes v2(r) ..... k .....

(rn/sec),

mac

a c (c for

(2.40)

and the period Tc can be determined by dividing the circumference by Equation (2.40) to obtain

/mo:

T~ = 2 7 r / ........ ~=-.

(2.41)

The angular fr~uency of the orbit is defined as 2_,~ = k ~ ~ wc = Tc

•

(2.42)

For an earth satellite, Equation (2.41) can be fu~her simplified. Since k G mere s= ms g Re,2 where m s is the mass of the satellite, m e is the mass of the earth, g is the gravitational con s t ~ t at the surface of the earth, and R e is the radius to the surface of the earth, k = 3.9865 × 1014 m s. Finally, Tc = 1.658668 X: 10 -4-ac3/2

(sec),

(2.43a)

and the circular velocity of the satellite ~ o m e s v2 =

2.3.2

3.9865 × 10 I4 ac

(mJsec).

(2.43b)

ELLIPTICAL ORBIT

It can be shown that the period of an elliptical orbit is given by Tc = 2rr ? ?........ '

(2.~)

2.4. SatellRe Coverage

41

where a e is the semirnajor axis of the ellipse (Figure 2.2). The angular frequency is

k

04 = . ~ a g ,

(2.45)

2.3.3 PA~BOLIC ORBIT The speed of a parabolic orbit is given by 2k Vp = ~n~l-'

(2.46)

where r is the radius of the parabolic orbit as shown in Figure 2.3. This equation is useful to calculate the velocity needed to escape from a circular orbit, The escape velocity is given by Yesc - X/2 vc,

(2.47)

where vc is given by Equation (2.40).

2.3.4

HYPERBOLIC ORBIT

The velocity along the hyperbolic asymptote (defined as r ~ ) Vhp -

. ..,

is found from (2.48)

where a is shown in Figure 2.3b, and the angular frequency is shown to be

%p =

2.4

.

( _ a ) 3.

(2.49)

Sate hte Coverage

The issue of satellite coverage is very impo~ant in satellite communications, navigation, weather forecasting, and surveillance. Some of the i m p o ~ n t issues

42

2. Astrodynamics of Satellites for Mobile Systems

in satellite coverage include the instantaneous coverage area as illustrated in Figure 2.5. The instantaneous area is represented by S(t), t because the coverage area is a function of time. The rate at which S ( t ) is viewed is given by d S ( t ) / d t . From the geometrical point of view, the instantaneous coverage area for a given minimum elevation angle ~ is illustrated in, F i b r e 2.6.

coverage

"4/ F i b r e 2.5

Instantaneous coverage area of a satellite.

Earth Coverage % 40 30 20 t0 L.

l ...........................

!. . . . . . . . . . . . . . . . . . . . . . . . . . . . .

1

1

2

............................. 2.5

Satellite Altitude Re

Figu~ 2.6 Percent of earth coverage for minimum elevation angle.

2.4. Satellite Coverage

43

The coverage area can be defined in terms of the intersection of a cone of hall-angle 0 that intersects the earth in a spherical cap. The half-angle of the cone 0 is 0 = cos-

~( cos~ ,,, i+--h-]R~J - ~'

(2.50)

where Re is the earth's radius, h is the height of the satellite above the earth, and 4~ is the minimum elevation within the region. The area of the instantaneous coverage is S(t) = 2 ~ R ~ (1 - cos 4~).

(2.51)

The portion of the earth's surface that is in view, Pvicw, is given by Pvicw = ~ (1 - cos ~),

(2.52)

which is plotted in Figure 2.6. Another representation of instantaneous coverage is shown in Figure 2.7. The average rate of change, of the surface swept out on the earth can be stated by

dS(t) 477"R2 ....... - ..........~ sin 0, dt T

(2.53)

where T is the satellite period and S(t) is the area swept during the time interval. The average time At for the coverage area to be refreshed follows from F~uations (2.51) and (2.53): T 1 - cos 0 At = . 2 sin 0

(2.54)

One of the most favorable orbits is that corresponding to polar, near-circular orbits. From Equation (2.51), coverage at high altitudes is greater than at lower altitudes. For near-polar, near-circular orbits, for a minimum elevation angle at the pole the inclination is 90 ° + ~, where ~ is given by Equation (2.50). ~ i s orbit configuration will have better coverage at the middle latitudes. For example, a satellite at 1 ~ km altitude and a minimum elevation at the pole of 1.0° will allow a range of inclinations from 68 ° to 112 °. Geosynchronous satellites, which occur when the semimajor axis is about 2 4 , ~ 0 miles, have a period of 1436 minutes, or 1 day. The main advantage of G E t is the large instantaneous coverage area, of about 42% of earth's surface. Latitudes as high as 80 ° can be se~iced. If the orbit has a small inclination, the satellite will show up tracing out a figure 8 as it moves from one side of the equator to the other.

44

2. Astrodynamics of Satellites for Mobile Systems

\ \

/ / /

\ \

h

\

O Figure 2,7 Instantaneous coverage geometry.

2.5

Some Terminology

In the diagrams shown as part of Figure 2.8, whether we are t-..alNng about a hyperbolic or elliptical orbit, the perifocal is the point on the. satellite orbit where the seconda~ is closest to the loci. The perifoca! distance is the. linear separation between the foci and the perifocus and is given by (a - c) = a (1 - e) for an elliptical orbit which has a semimajor axis a and e~centficity e. The perihelion is the closest approach to the sun; the perigee is the closest approach to the earth, Perihelion and perifocus are measured from the center of mass, but perigee height is measured from the surface of the earth. ~ i s terminology is useful because we're often interested in the height above the surface for low~.altimde satellites. In elliptical orbits,, the most dis~nt point from the loci is called the apogee (c + a). The straight line connecting the apogee, the. perigee, and the two loci

2.5.

Some

Terminology

45

orbit above planeof descendingnode

J............ 5~..,._ .....

c

perifocal ~ "xxxdxi~,ance'

line of

orbit beIowplane

"',~

apsides

Figure 2.8 Terminology for elliptical orbit.

is called the line of apsides. If dp, d~, and Re are. the perigee height, apogee height, and radius of the earth, respectively, then for a low earth satellite,

a = R ~ +~(dp + d.).

(2.55)

To define an orbit we need to identify size, shape, and inclination. The inclination 0i is the angle between the orbit plane and a reference plane, which for the earth, is the equatorial plane. The intersection of the orbit plane and the equatorial plane through the center of the earth is called the lines of nodes. For an earth satellite, the ascending node is the point in its orbit where a satellite crosses the equatorial plane going from south to north. The descending node is the point where it crosses the equatorial plane from north to south, as shown in Figure. 2.9. Finally, there's a simple method to specify where a satellite is in its orbit. The true anomaly fl is the angle measured at the loci between the perigee point and the satellite. The mean anomaly M is 360 (At/~ degrees, where T is the orbital period and At is the time since perigee passage of the satellite. Thus, for a satellite in a circular orbit M = fl, the eccentric anomaly Ois. the angle, measured at the center of the orbit between perigee and the projection of the satellite onto a circular orbit with the same semimajor axis as shown in Figure 2.1.0.

46

2. Astrodynamics of Satellites for Mobile Systems

°°"°e

(, perigee ~

.,- "~"

e0uto,

) plane of earth "" ascendingnode

Figure 2.9 Orbital elements in satellite motion.

Location of Satellite

t

I ............................ prim"'a~

J

Perifocus

Figure 2.10 Definition of eccentric anomaly,

2.5. Some Te~inology

47

The mean and ezcentfic anomaly am related by M=

~-

esin O,

where e is the eccentricity. O is then related to ~ by tank

\1 -

e}

tan(0/ 2).

(2.56)

Chapter 3

Attitude Control and Navigation

3.0 Introduction to Attitude Control in Satellites Control of the attitude of a satellite is important to increase its usefulness and effectiveness for establishing reliable communications links. If a satellite is stabi~ lized in orbit, its directional antennas can be properly pointed toward earth, solar ~ a y s can be properly pointed pe~endicular to. the sun's light (any deviations from this would result in less generated power), satellite sensors can be pointed properly in the celestial space (star, the sun, etc.), and transfer orbits, can be executed for tracking purposes and to properly align the satellite centerline for futures firing of motors. Finally, a satellite is exposed to a series of adverse torques generated by atmospheric drag, solar wind, magnetic fields, solar and space radiation gravitational fields from multiple celestial bodies, and micromete~ ofite (space junk.) impacts. Proper attitude control of the spacecraft will address such adverse t~ques. If a satellite is to be put into low earth orbit (LEO) and is to be launched directly into such an orbit, then most likely the satellite will have a low spin rate.. Once in orbit, its attitude can be changed by incre~ing the spin rate with a different orientation of the spin axis, or it. can be three-axis stabilized, in which case the antennas and solar arrays will be deployed only after obtaining a zero spin rate. Such desired attitude will be achieved by the use of reaction (momentum) wheels, magnetic torques, and thrusters. If a satellite is to be put into geostationary orbit (GEO), then the satellite is first launched into a low elliptical orbit which is later circularized by firing, a series of thrusters. Then the satellite is put into a much higher but highly elliptical orbit with its apogee altitude equal to its final satellite orbit altitude (about 24,000 miles). By a sequence of firing the apogee kick motor at apogee, the satellite orbit is circularized. Once the satellite is in. the desired orbit, the satellite orbit is changed into its desired final attitude. The attitude and ~iculation control subsystem (AACS)in a satellite performs both functions of orbit and attitude, control. Figure. 3.1 shows the pitch, yaw, and roll coordinates of a satellite in orbit. Most communications satellites are spin-stabiliz~ systems. The satellite is spun ~ound a symmetrical axis. The momentum imparted by the spin tends to 48

3.0. Introduction to Attitude Control in Satellites

x..._

49

Yaw ""~-'~~-

........................ -4

Roll ~' Pitch

Figure 3,1

Definition of pitch., yaw, and rolL.coordinates.

keep the axis of spin fixed in space. Simultaneously, the antenna is mechanically despun against the rotation of the outer body so that it remains fixed toward a region of earth. Antennas located on the spin axis will cause a doughnut-shaped beam to remain in a f i x ~ position relative to the earth. In spin-stabilized satellites, the AACS maintains the satellite in the desired attitude and orbital position while the satellite is subjected to a variety of upsetting toNues. Ngure 3.2 shows an overall block diagram of a typical AACS system onboard a satellite. A variety of sensors gather data from celestial stimuli: and process that information internally, The command and. data handling subsystem (C&DH) serves as the processor of internal commands generated ~ the result of dam recorded from the AACS sensors. The result of command and data processing will activate a series of actuators (driven from their drive electronics) which control the attitude actuators. Command and telemetry signals are also processed in parallel. Among the earth sensors of the AACS subsystem~ the following are most common: the reaction wheel assembly (RWA), the inertial measurement unit (IMU), the sun sensors, the celestial star assembly (CSA), and horizon sensors. In Figure 3.2 we find the AACS-sensor interaction block diagram. The CSA provides the C&DH computer with precise timing of a s~r image crossing at a detector slit, The C&DH computer matches the sensor data against a resident star catalog, thus determining the attitude of the sensor with respect to the inegiaI space, Given the known mounting geometry of the CSA with respect to the satellite, the inertial attitude of any satellite object can then be calculated via a straightfo..~ard coordinate transformation.

3. Attitude Control and Navigation

50 External Stimiii Stars

Subsystem Interfaces

1 I,

{ ...........

1

............t.......

tValve C o n t r o ~

C&DH

'....................................... "

Subsystem

-'-~, R-'t--'-

By Body Accel.

i

'~v'u IMU

=T h [

................

Engines

#n/off C o n t r o l [ C a t - B e d

i...................................L , ~ ~ i

Control i n t e r f a c e s

,

I

IL

....................

"

" q ..............Posffion

Htr

I Ant. GDE

31

1

. . . .

, ...........

........:.........-.........................

]

I I i Sun

i

I

[~SID~H ............................ !

~ 1 1

[ -S-8~q

- P,.,,r,,

1 Cm~.

~.............~

..... .

~

' - - 1 i

Array GDE

i Pos.ion ~ S o ! a r

.... c

.....................

|

..................................

~LM

._

,

...........................

AACS ,~ .

~ottware

;;.;:

I

1I

I Torque

_ _ t ........ 1 ...........

...............

- ....................... = . . . ....................... .......

. . . . . . . . . . . . . . . . . . . . . . . . . . .

_ =.J ........................

............................

I

GDE.=Gimbal Drive Electronics SSD=Sun Sensor Detector | SSE= Sun Sensor Electronics

I

.............................

['i

L

F i b r e 3.2 Typical block diagram, of a satellite AACS.

3.1

Physical Principles of Sun Sensors

The sun sensors t r a n s f o ~ the light energy from the sun into an electrical impulse or current which, through a transimpedance netv~ork, is converted into a voltage signal. The voltage is amplified and the measurement is used ~ a way of tracking the sun's presence. The sun sensors are needed to orient the spacecraft toward the sun; doing so will allow the attitude algorithms to orient the spacecraft for maximum power acquisition using the solar arrays. The electronics of sun sensors is v e ~ simple, since the sun can be considered only as a point source which is bright enough not to. be confused with other stellar bodies. Sun sensor measurements are needed for thrusters firing in order to position the solar w a y s and whenever phase-angle info~ation is needed. There are basically three kinds of sun sensors: analog sensors, sun detectors (to protect instrumentation), and digital sensors. The digital sensor is widely used for spinning spacecraft. A typical sensor is shown in Figure 3.3. For full 47r degree coverage, up to four sensors can be counted with overlapping fields of view. As illustrated in Figure 3.4, the sunlight is diffracted as it enters the sun sensor and illuminates a slit pattern. The slits, which underneath have corresponding photocells, are divided into several rows. Four classes of rows are usually present: (1) threshold adjust,

3.1. Physical Principles of Sun Sensors

51

Sensor',) f # ~ ~-~"'-,-~

~N.~

..................

iElectrOnics !

Figure 3.3 Basic sun-sensor configuration for 4xr coverage.

(2) sign bits, (3) encoded bits, and (4) inte~olation bits. The threshold adjust slit is half the width from any other fully illuminated slit photocells as long as the sun image is narrower than any inte~olation slit. A bit is " o n " if its photocell voltage is greater than the threshold adjust photocell voltage. Therefore, " o n " means that an interpolation slit is more than half illuminated. The sign bit determines which side of the sensor the sun. is on. The encoded bits provide a discrete measurement of the linear displacement of the sun image relative to the sensor center line. Either a Gray code or a binary code is used to convert the illuminated slits into a digital pattern. Gray code is the most. commonly used encoding method.. From the digital pattern the sun angle 0 is calculated (notice that !o = I cos ~, where I is the current illuminated by the photocell. 3.1.1

ELECTRICAL FUNCTIONAL BLOCK DIAGRAM

The digital, functional block diagram is shown in Figure 3.4. The sun. sensor consists of several two-axis digital-type photocell detectors, and this generates

52

3. Attitude Control and Navigation

Sun

/

/1

t

Sun

TOP View Reticle Slit Pattern Interpolation Bits

]1

l]'q

r . . . . . . . . . . !.......... 1.......... |

ilrGray Coae

t

_~

i Bits(,loo,.)]

[

Photoc~.!ls

Bits(l~ooo111...) } V

Figure 3.4 Details of digital sun-sensor components. the input signals to the sensor electronics. The detectors are used for 47r steradian coverage and sun angle data. Each detector generates solar aspect data and automatic threshold adjustment signals. The sensor power is generated from the satellite internal power bus structure. As shown in the electronic block diagram of Figure 3.5, sunlight hits several phot~ell detectors, depending on their location and the sun angle. Different currents are generated, each co~esponding to a different bit of a word. The bit pattern is sent to a sensor circuit. There can be several sensor circuits (e.g., two sensors in Figure 3.5), and each sensor generates its own bit pattern. For a twoaxis sun sensor (e.g., x- and y-axes), sensors which are located, on both axes will generate a bit p atmm. In the example of Figure 3.5, the two sensors for each axis will each generate a bit pattern. A bit comparator is used to compare the signal strength of each bit pattern, and data is output on a shift register. A companion series of circuits selects which sensors will "be used for obtaining the data, depending on the observed strengths of their signals.

3.1.2

NOISE PROBLEMS IN SUN-SENSOR CIRCUITS

One of the most sensitive components in the sun-sensor circuit of Figure 3.5 is the comparator circuits and data-good logic circuits. ~ e s e circuits suffer a lot

3A. Physical Principles of Sun Sensors

53

Drivers

o

..................................l ........

commands .

Sensor 2

[

t

Star CMD.

Clock Data

Figure 3.8 Electronic functional block diagram of a digital sun sensor.

from oscillation problems. The oscillation problem most often has more to do with the design of the circuit than with the operational amplifier, One reason for oscillation is that dynamics associated with load applied (shift. register in this case) or the feedback network connected around it combine with the openloop transfer function of the amplifier to produce feedback instabilities~ Proper compensation is essential :for achieving optimum per[ormance from virtually any sophisticated feedback system. If stability is the only concern, lowering looptransmission magnitude usually is sufficient for systems that do not have rightha!f-plane poles in their loop transmission~ Better compensation is required when high desensitivity over an extended bandwidth or wideband frequency response is desired. Another common cause for oscillation is excessive power connection impedance. Comparators have a lot of voltage gain and quite a bit of phase shift at high frequencies; hence, oscillation is always a possibility. Most comparator problems involve oscillations. This problem is worst at high frequencies because of the inductance of the leads that couple the power supply to the operational amplifier. In order to minimize woblems, it is impor~nt to properly decouple or bypass all power supply leads to amplifiers without internal decoupling networks. Good design practice will use large values (> 1 /A~') solid-tantalum electrolytic

54

3. Attitude Control and Navigation

capacitors from the positive and negative power supply to ground on each circuit board. Individual amplifiers should have ceramic capacitors (0.01 to 0.I /zF) connected directly from their supply terminals to a common ground point. The single ground connection between the two decoup!ing capacitors should also serve as the tie point for the common input signal, if possible. Because of series inductance, lead length on bo~ the supply voltage and the ground side of these capacitors is critical. If low supply currents are anticipated in sun sensors, crosstalk between operational amplifiers can be reduced by including small series resistors in each decoup!ing network, as shown in Figure 3.6. The open loop transfer functions of many operational amplifiers are dependent on the impedance which is connected to the noninverting input of the amplifier. If a large resistor is connected in series with this terminal, the bandwidth of the amplifier could diminish, and this will lead to oscillations. In these cases, a capacitor should be used to shunt the noninve~ing input of the amplifier to the common input signal and power supply decoupling point.

3.1.2.1 Grounding Improper grounding is a frequent cause of poor operational amplifier performance. One frequent grounding problem is the result of voltage drops in ground lines 0.1 pF ---"I | .............................................. [ ....................... --->.v'X/% ........................................................................................................ ~-~.i.~.............................. I

.I.

WTo Input Signal Common Tie l ............................ II 0.1 p.F

Une impedance = R ..................

~-,/x,,,'X,

I

~J-

Figure 3.6 Decoupling network for operational amplifiers.

3.2. Reaction Wheels: Physical Principles

55

R2 Rt

Vo

Figure 3.7 Improper grounding of operational amplifiers.

as a result of the current flow through these lines, known, as ground impedance coupling, In Figure 3.7, both. the signal and power supply are connected to the same single ground point. However, a potential Vg is created when the current through the load also sets a potential on the noninverting input of the operational amplifier, Now the amplifier output voltage with respect to. the system ground is given by

V°

Rf = -R~

Vi

+

Rf + R i RI

Vg.

(.3.1)

The error term from. Equation (3.1) can be significant, since narrow printed circuit board conductors and connector pins can have considerable impedance (resistance and inductance.).

3.2

Reaction Wheels: Physical Principles

Momentum and reaction wheels are. used for the storage of angular momentum. They have several purposes in a satellite AACS: first, to add stability against distorting toNues; second, m absorb cyclic torques; and third, m transfer momentum to the satellite body for the execution of slewing maneuvers, These devices depend on the momentum of a spring wheel, h = Ioz where 1 is the moment of ine~ia about the rotation, axis of the. wheel and ~ is the angular velocity.

56

3. Attitude Control and Navigation

Figure 3.8 shows a diagram of the typical reaction wheel. In general terms, a momentum or reaction wheel consists of a housing that. contains the rotating wheel, bearing assembly, drive motor, and drive/control electronics. The motor in the momentum or reaction wheel produces a net acceleration torque about the wheel (angular velocity which is constantly changing direction produces a torque). The drive/control electronics provides two modes of operation: (I) constant anguI~ momentum by providing a constant speed, and (2) torque control motion. In constant angular speed, the momentum is maintained at a desired value. In the torque control mode, the wheel speed is changed using a close feedback system to respond m external torques, with the objective of mai.nt~Jning a determined direction for a pa~icular axis. ~ e r e are two types of motors: AC two-phase induction motors and DC brushless motors. AC motors do not require brushes or slip rings; therefore, they have higher reliability and a longer life. However, £ne efficiency is low since they have a low torque and need a high operating speed. DC motors, however, can provide high to.rque at low speed. The n o d a l brash comrp:atators are replaced with a type of electronic commutation.. From a reliability point of view, the lubrication system in reaction or momentum wheels must be well controlled: when ~ e seal is exposed m the space environment, it tends to evaporate somewhat, increasing the friction, of the bearing

Thin Cover FlatWeb of

\\~t ..........

Shaft ~ _ / B.earing

Hyw

........

~zz:/L17z:

_.i LZ3,,L,L ............................................... ~~"

~i

~ 2 ~ ..... ;.____i"ziiiiiz 2_iiii 2....

L..~rzzi z : z z . 2 z z

KN _..............N k-W jHoo og

of....Flywheel"~-~¢&~v.;~

x ", .i~.i ....i _~i..:~.'i~:~t~ ~ ' # ~

!i

• %%

Mot

es

/

Thin Cover

!

Bearing

~

"

.............

~'~-~"

Figure 3.8 Reaction wheel assembly.

Hermetic

Connector

Powerand Control Circuits for Motor

3.2. Reaction Wheels: Physical Principles

57

to a point where the drive/control electronics can be damaged. Low-vaporpressure lubricants and even dry lubricants are cu~ently used for bearing lubrication. In essence, momentum and reaction wheels are like gyroscopic actuators m offset disturbing toNues on. communication satellites. For example, in order t.o properly point a recently deployed anmnna (the deployment itself generates a disturbing torque), the momentum or reaction wheel is used m compensate for the disturbing torque. Normally, three reaction wheels are used to control a satellite, with the wheel axes aligned with the body principal axes; a redundant fourth wheel is often used in case of a failure, of one of the main wheels. The momentum or reaction wheels are powered off during launch. As soon the satellite is put into orbit, the wheel is spun. ~ e equivalent generated torque is used to despin the: satellite (most satellites are released with a spin imparted by the launch vehicle). Though thrusters can also be used in attitude control (e~g., despin), they are used only if (1) a momentum or reaction wheel has failed for a particular axis and (2:) the disturbing torques exceed the control of momentum wheels. Therefore, reaction/momentum wheels also play a real role in decreasing the fuel consumption of thrusters.

3.2.1

REACTION WHEEL FUNCTIONAL BLOCK DIAGRAM

There are three main components in the design of a reaction whe~l: the motors/ bearing unit, the wheel, and the drive/control electronics.. In the. motors/bearing unit, the pe~ane.nt magnets of the rotor provided a desired magnetic field. The stator consists of three-phase windings wound on sheets of a metal core. Commutation is accomplished electronically. The associ~ ated electronics is found within the wheel housing; status signals concerning speed and speed direction are also found. Precision bemngs provide long life if lubricated properly. The lubricant will remain within the bearings if migration or outgassing is prevented by seals. For a particular angular momentum desired the wheel mass has to be optimized so as to keep the motor power consumption within specified limits. Generally, wheels that resemble: either a disk or spoked wheel, have been designed. A simplified block diagram of a typical drive/control electronics assembly for a reaction wheel is shown in Figure. 3.9. The drive/control electronics is the interface between the motor/bearing unit and the wheel. The drive/control assembly receives commands from the C&DH subsystem and determines the torques, speed, and speed direction of the wheels. The electronics are normally equipped with up to three channels for satellites equipped with up to three wheels. Most

58

3. Attitude Control and Navigation

torque commanc!~

from Motor I- -iLH~at! . Comparator~I

|

~

t t

1 !

current TLM

t

Hall Generators

...........................

]

Tachomelor l L I(_F .............................!

Figure 3.9 Reaction wheel drive electronics~ reaction wheels have the DC-DC converter to supply the needed voltages if the input power is unregulated. Torque commands that originate in the C&DH subsystem are used to generate a pulse-width modulated signal which will drive each of the three phases of the motor windings, A series of FET drivers will do the job of providing the motor driver currents. The block diagram in Figure 3.9 also shows a current sense circuit to be read by the C&DH software as an indication that the driver circuits for the three-phase motor are working properly; current TLM circuits; and tachometer sensing and TLM circuits. Such circuits provide a measure, of how fast the wheel is rotating. Notice from the figure that a series of Hall sensors (at each motor winding) together with Hall generators/ comparators s e r e as the feedback me~hani.sm to the motor drive electronics by providing the proper enable signals.

3.2.2

NOISE PROBLEMS IN REACTION WHEEL ASSEMBLY

~ e most impo~ant noise issue found in reaction wheel electronics is the noise associated with motors, power conve~rs, and switching transistors. The noise provided by such. sources far outweighs the possible sources of interference from

3.2. Reaction Wheels: Physical Principles

59

other major components such as comparators and digital logic. A detailed study of noise problems in power converters will be addressed in a later section, here we cover mainly interference concerns due. to DC motors and switching devices. From the DC source point of view (the DC source is provided by one of the outputs of a DC-DC converter inside the reaction wheel drive electronics), the DC motor behaves like a variable but unidirectional emf E behind a resistance R. The latter consists of an "external" part (.the cu~ent-limiting starting resistor) and. an "internal" part (contact resistance between fixed and moving parts), plus conductor resistance or armature resistance. A DC motor is shown in Figure 3.10a. An equivalent circuit of the motor is shown in Figure 3.10b. The motor is fed from an armature voltage source supplying a terminal voltage '~. The field coils are separately fed from an excitation on field voltage Vf. The total resistance of the armature winding, including the blush contact resistance, is lumped into the armature resistance R ~. The difference between Vaand the generated emf E is equal to the voltage drop across the armature, resistance, that is,

'~

-

E

=

(,3,2)

RaI..

ta

Va Tm

(a) Power Flow

DC Source

LOAD Power Flow

(b)

Figure 3.10 (a) DC motor representation, (b) circuit representation of motor.

3. Attitude Control and Navigation

The motor emf E is given by E = Kwh,

(3.3)

where K=

p = n = a = o~= =

pn 2 ~a

number of poles in motor number of conductors in armature windings number of paralleled paths in armature windings angular velocity (rad/sec)of windings uniform magnetic core flux density (Wb).

The motor torque is given by T = K~b!a.

(3.4)

In the. DC motor, as the cu~ent to the rotor windings is connected to and disconnected from the DC source through the commutator segments, arcing at the brashes occurs due to the periodic interruption of the current in the rotor windings (inductors). The arcing has a very high-frequency spectral content. The arcing spectral content tends to create radiated emissions in the 2.~ MHz to 1 GHz frequency range. In order m suppress the arcing, resistors and capacitors are placed across the commutator segments. These can be implemented to the f o ~ of capacitor or resistor tings attached directly to the commutator or resistive rings placed around the commutator. Another source of high-frequency nNse and associated radiated and conducted emissions is that produced by the switching electronics of the driver transistors shown in Figure 3.t l. ~ e s e driver circuits are used. to change the dir~tion of rotation to provide optimum position control of the DC motor, A typical drive circuit is shown in Figure 3.1ia. When transistors T I and T4 are turned on., current flows through the commutator and the rotor windings, causing the rotor to turn in one direction (winding #1). "When these are tuxned off and transistors T2 and T3 are tumext on, the rotor turns due to windings #2. These driver circuits ~ e usually connected to the motor via by pair of wires as shown in Figure 3.1 lb. Because of t h e ~ a l cooling, the motor housing is usually attached to the metallic frame of the product on the heat sink. This causes a large parasitic capacitance Cm between the motor housing and reaction wheel assembly frame. This capacitance

3.2. Reaction Wheels: Physical Principles

61

:?v.c ................................

T3

[

LB

•

~, ..............."....................................... ... , ~

Lc

Lo

T2

(a)

Common Mode Choke

(b)

[ F i b r e 3.11

kooPArea

"

~ . .

(a) Motor drive transistors, (b) conversion of common-mode: current into differential-mode cu~Tent,

62

3. Attitude Control and Navigation

provides a path for common-mode currents to pass through the connection wires from the rotor to the motor frame via capacitance between windings and eventually to the reaction wheel frame. The current provided to the motor by the driving transistors has fast rise time spikes due to the constant switching, and so does the current through the commutator. These spikes have very high-frequency spectral content, and can couple to other parts of the reaction wheel drive electronics and then radiate. The radiation potential tends to be a direct function of the loop area occupied by that current; the lm~gerthe loop area, the larger the radiated emissions. In order to block this common-mode current, a common-mode choke may need to be placed in. the driver leads, as illustrated in Figure 3.1 lb. The capacitance Cm is also represented in Figure 3.1 l b. Concerning the switching of transistors, Figure 3.12 shows the typical voltages and current waveforms. The switching time is shown as t, the delay time is ta. The figure shows the waveforms for gate-source voltage Vgs mad drain current ID of the power transistor for switching off. The drain current/~ continues to flow until time ts, called storage time. During this time inte~al, the charge carriers will be removed from the depletion region. After the storage time, the collector current falls to zero. The fall time of the collector current is rather short, usually in the range of I0/xs to t 00 ~ , depending on the rated power of the transistor. The Fourier transform of Figure 3.12a waveforms will show emission levels of noise covering a wide spectrum as shown in Figure 3,12b.

.................................................................................

...............................

............

I D

I .................................................

!..

..............

,o/,,.... i v

.

.

.

.

.

.

~

i . .......................

,

~

Figure 3.12 (a) Power transistor switching process, (b) s~ctrum of switching process.

3.3. Intrinsic Noise in Operational Amplifiers

3.3

63

Intrinsic Noise in Operational Amplifiers

Since operational amplifiers are composed of active devices (transistors, FETs) and resistors, operational amplifiers experience shot (Schottky), thermal (Johnson), and flicker(l/f) noise in an intrinsic manner. Undesired electrical signal present in the output voltage of an operational amplifier is classified as noise. The internal operational amplifier noise is modeled simply by a noise voltage source V,~. As shown in Figure 3.13, Vn is placed in series with the noninverting input. On data sheets, noise voltage is specified in microvolts (RMS) for different values of source resistance over a given fr~uency range. For example, the 74I operational amplifier has 2 #V of total noise over the frequency range of 10 Hz to 10 kHz for source resistor R i between 100 ohms and 20 kohms. The noise goes up in direct proportion to R i a s R i exceeds 20 kohms. The noise current in the operational amplifier input appears as a bias current. The output noise voltage from this bias current becomes (3.5)

Vnl : inR r.

Of Rf

VO = - Vn ( t + R f / R i )

~V

<

<<, Rf It Ri

~7 Figure 3,13 Internal noise modeling of an op~amp.

64

3. Attitude Control and Navigation

The noise voltage Vn appears on the operational amplifier input as an AC "offset"' voltage and thus appears on the output as

Vnv = Vn( R~. + 1 ) , \,Ri

(3.6)

where

to:

(Schouky noise)

q = 1.6 × 10 -I9 coulombs (charge of one electron) ~c = average current in semiconductor BW = frequency bandwidth (Hz) over which the noise is considered .

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

Vn = "v/-4K TR(BW)

(thermal noise)

K = 1.38 × 10 ~:23 joule/K (Boltzmann constant) r = temperature in kelvins (degrees Celsius + 273) R = resistance (wire-wound or metal-film resistors) The total output voltage due to noise becomes

v~o = x / v ~ + v :2~,,.

(3.7)

Recommendations to minimize internal noise include the following: 1. Do not connect a capacitor across the input resistor or from the inverting ( - ) input to ground. The R iC combination will have a lower irnpedance at higher noise frequencies than R i alone, and gain will increase with frequency and make the situation worse. There will always be a few picofarads of stray capacitance from ( - ) input to ground due to wiring. 2. Do connect a small capacitor (---3 pF) across the feedback resistor. ~ i s reduces the noise at high frequencies. 3. If possible, avoid large values of R f and R i.

3.4 Star Camera: Physical Principles In addition to the sun sensor previously discussed, there are other sensors that help orient spacecraft relative to the earth. One such sensor is the horizon sensor, the principal means for directly determining the orientation of a spacecraft with

3.4. Star Camera: Physical Principles

65

respect to the earth. The other widely known sensor is the star sensor, also known as the star camera. Star cameras measure star coordinates in the satellite's reference frame and provide altitude information when. the observed coordinates of the. measured stars are compared with known data obtained from a star catalog. Star sensors are the most accurate of attitude optical sensors. However, they are usually heavier and more expensive than other sensors. Furthermore, computer software requirements are extensive, because measurements can be preprocessed and are then identified against the catalog. Star cameras may have some problems because of occultation from the sun and the earth. The elements of a star camera are shown in Figure 3.14: a sun shield,, an optical system, an image definition device that defines the region of the field of view visible to the detector, the detector, and an electronics assembly. The sun shade is usually employed m minimize exposure of the optical system to sunlight and the light scattered by space dust particles. Star cameras are not useful when the sun is within 30 ° m 60 °. The optical system of a star camera is made up first of a lens which projects the image of a star field onto the focal plane. The image definition, device selects a portion of the star field image in the camera's field of view which, will then be visible to the detector. The detector ~ansforms (Figure 3.15) the optical signal into an electrical signal. The most frequently used detector is a charge coupled device (CCD). The electronic assembly filters then amplify the signal received from the CCD. The. main component, of the star camera is the CCD array, which transforms the weak light from stars into small electrical signals. Figure 3.15 shows timing signals and clocks needed by the CCD array. The output analog of the CCD is put. through a series of feedback amplifiers which will strengthen the CCD output signal. The amplifiers are commanded and controlled by another

Image Definition Photo-detector

Optical System

I

Electronics Deflecting Coils

Figure 3.14 Elements of a star camera.

"~n Shieid

3. Attitude Control and Navigation

Input ........................... "ower ~,,-15/+15V :"on~ver~te,s- ' J ~~wu[~- } -~- ~ F ' - 0v .

......

.

.

.

.

.

.

.

.

AmpL# i

'................................ [ / V D I , Dig,ta,Outputs i Cony.~_~ Video Output

AmpL#2 Ampl"i~;~

CCD

Timingcontrol CLKDrivers ControlCLKs

Controls

LineSync. E TimingControl: ASIC

CMDs

ControtAStC .

.

.

.

.

.

.

.

.

.

Figure 3.1.5 El~tronic functional block diagram of satellite, star camera. FPGA whose output digital signals provide gain-select and offset inputs to the amplifiers. The amplified analog outputs are digitized in an A/D converter, and the digitized data is processed using a DSP FPGA. Output drivers will transmit the output digital data to the inertial measurement unit (IMU) for attitude control. Interface transceivers process command data from the C&DH and stares words are read back by the same software. 3.4.1

NOISE CONCERNS IN STAR CAMERA E ~ C T R O N I C S

We next examine the noise issues that may arise from analog circuits in star cameras, although interference concerns may exist in. digital electronics: clock signals to the CCD and DSP chips, as well as noise in analog-to-digital converters. The major interference concerns here lie in the coupling of noise produced by digital electronics into sensitive analog circuits. "We first, examine some simple noise coupling scenarios in analog circuits, and then we explore the noise sources from digital circuits.

3.5

Noise in Amplifier Circuits

A simple amplifier circuit m illustrate the effect of noise in analog circuits is shown in the circuit of Figure 3.16.. As is well kmown in amplifier circuits, the

3.5. Noise in Amplifier Circuits ,

I= '~/R i

(~)

v,

",,/'VX,

...................................................................

Rf Ri ....................................................[.."..~ ..... ~ +V

[

v

67

1+....-f" ...........~ '~-v

Io=IL+I

~-

Vo =- Vr Rf/Ri

<

Figure 3.16 Simple amplifier circuit,

voltage Vd between ( + ) and ( - ) (i.e,, the noninverfing and inverting inputs) is practically zero. The current drawn by any of inputs is negligible. The current t through R i is found from Ohm's law, I = V~,

Ri

(3.8)

where R i also includes the source resistance. The voltage across R f is given by

Rf

Vet = R-]i ~"

(3.9)

The output voltage on the load is given by Vo = -R~--2"~, Ri

(3.10)

The closed loop gain of the amplifier is given by ~, Rf A = ~ - ---. Ri

(3.1I)

The minus sign simply shows that the polarity of the output VL is invert~ with respect to ~. The load current I L is 1L

= VL---RL.

(3.1.2)

68

3. Attitude Control and Navigation

The total current in the output is given by adding Equations (3.8) and (13.12): IL + L

!ore-

(3.13)

We now include the noise source Vr, as shown in Figure 3.17. The output voltage seen at the load due to the noise is given by '¢'L. =

1 + Ri

V..

(3.14,)

The load currently in the output due to the noise is given by IL n =

VLn.

(3.15)

RE

The total current in the output is given by adding ~ e previous equations" lout (tota~ = ILn + L-ut-

(3.16)

The total output voltage seen by the load is Vout (total)=

-~/

Vi +

I +

Vn.

(3.17)

If the noise source were to act on the inverting ( - ) input, then ~ u a t i o n (3.17) becomes Eut(total) =

-

F~

!=VilRi

~--"v%'%., •

E-

Ri

OV

¥

---------

vi

1 +

Vn.

(3.18)

+V = - V i Rf I Ri + (1+ Rf I Ri) Vn

@

vn

V

Figure 3.17 Example of inverting amplifier in the presence of noise.•

3.6. Noise Sources 3.6

69

Noise Sources

Highly sensitive analog circuits are susceptible to RF noise generated by nearby digital circuits. Clock waveforms are the main source of interference in digital circuits. We now discuss some of the dining and spectrum fundamentals of clock waveforms and their noise potential. Clock waveforms are periodic trains of trapezoidal pulses as shown in Figure 3.18. Each clock pulse has an amplitude of V (volts), a rise time given by tr (nsec), and a fall time given by tf (nsec). A pulse width T(#sec) is measured between the 50% points (V/2) of the w a v e f o ~ amplitude. The Fourier transform of the pulse train shown in Figure 3.i8 is, in magnitude form (i.e., magnitude of harmonics),

_

2V7"(sin(n.~r/T)'~

(sin(nTL~TT)),

(3.19)

where n ¢ 0 and tr = tf (usually a valid approximation). If we replace the discrete spectrum of Equation (3.19) with a continuous spectrum by letting f = (n/~, we obtain Envelope of

T ~,, ~rf ] "\ ¢:'trf]"

(3.20)

The "bounds" of the. spectrum of Equation (3.20) are shown in Figure. 3.1.9. From the spectral bounds, it can be obse~ed that the high-frequency content of

v(t)

tr

tf

T

Figure 3,18 Trapezoidal clock signal.

3. Attitude Control and Navigation

i, v

2V-tiT

..................................................................................... 1~'~.~__.._22 0 d B / d e c a d e

v n"

-40 d B / d e c a d e

i ,

1/~

i I

flHz)

l/~r

F~a.re 3.19 Sample spectrum of trapezoidal clock signal..

a clock signal is due. primm!y to the rise and fall times of the clock pulse. Pulses with small rise and fall times have a larger high-frequency spectrum than pulses with larger rise and fall times. Two important facts that can be deduced from the potential of clock signals to serve ~ noise sources: 1. Clock signals can produce noise levels over a large, frequency spectrum as shown in Figure 3.19. 2. ~ e magnitude and frequency of such noise levels is dependent primarily on the rise time (t r) and magnitude of the clock pulse, as shown in Figure 3.19,

3.7

Simple Electromagnetic Noise Coupling

"We now consider the scenario of a digital circuit in the close presence of an analog circuit, as shown in Figure 3.20. In the figure, L is the length of the susceptible circuit, s is the height of the susceptible circuit above the ground plane. It is assumed that the noise source and susceptible circuits are coplanar. Furthermore, we are assuming in this simple model that L << 3. (electrically short circuits) where A = c!f; f is the highest harmonic of the oscillator circuit of Figure 3.20 which falls within the spectrum of Figure 3.19. Notice also from

3.7. Simple Electromagnetic Noise Coupling Noise Source Circuit

PCB Trace 100 MHz OSC

i5V

~

71

Etectromagneti ,4, Fields

Hn i

~d V

Vs

I I I

l

.................

Vn s

In Area- A = L x s

Susceptible~ Circuit

";RL 1

Figure 3.20 Noise coupling of digital circuit into analog circuit.

Figure 3.20 that A = sL, where A is the area of the receiver loop. The noise and susceptible circuit are separated physically by a distance d. When such assumptions are valid, the terms Vn and I~ in. Figure 3.20 can be represented by Vn = jo~/z,~H~

(3.2 i)

In -

(3.22)

-jwCLsE

I,

where HI~ and El. are the incident magnetic and electric fields, which are n o d a l and tangential, respectively, to the plane of the susceptible circuit as shown in. Figure. 3.20. The term w = 2~r~ where f is the harmonic frequency of interest of the oscillator; P~o is the magnetic susceptibility (4~ X 10 -7 H/m); and C is the per-unit-length capacitance of two parallel lines (assume the "noisy" and "susceptible" circuits in Figure 3.20 are approximately in parallel): C =

~°er ,°(s)

(in F/m).

(3.23)

",,r ~, /

r~o is the equivalent radius of printed circuit board (PCB) cross sections, e r is the relative permeability of the surrounding medium (~sumed homogeneous and nonfe.lvomagnetic), eo is 1/36~ × 10 -9 (free space permittivity).

72

3. Attitude Control and Navigation

For wire circuits behaving as antennas (i.e,, the oscillator circuit in Figure 3.20), because they radiate, the far-field approximation is given by dfar.field ~;> 3A. We will use the computation of far fields using a two-conductor line scenario (similar to an array of elements). The fields generated by each of the lines are superimposed. The radiating model to be used is that of a Hertzian dipole. The total ]E 11max will be that resulting from differential-mode current ID and common-mode current Ic.

3.8

Common-Mode and Differential-Mode Currents

To better understand the nature of both differential-mode and common-mode currents, Figure 3.21 is worth dwelling on. Differential mode currents ID are currents that, under ideal conditions, exist in every current-c~ying circuit. They axe the currents generated by the voltage source in F i b r e 3.21. Notice that in the circuit of the figure, the cu~ent i D goes in opposite directions as it travels the closed loop of the circuit. The fields

Hn i

15V

i

t t

E1C

E i max = E i net (diff) + E i net (comm)

td I !

!

I

I

1

i E2C

~ e t ( diff )

T Enext(comm)

Figure 3.21 Illustration of common-mode and differential-mode cmTent.

3.8. Common-Mode and Differential.Mode CurrenN

73

generated by each current I D, EID, and E2D, are in opposite directions and only a net field Enet(diff) survives. The fields do not cancel each other out because even though we have the same current l D, the forward and return paths of the currents are hardly ever symmetrical.; therefore, each current path generates its unique field. Common-mode currents (lc)are the "parasitic" currents that become part of a circuit after they "couple" into the circuits from external sources (e.g., other nearby circuits using the same ground plane) through parasitic components (e.g., parasitic capacitances m ground in Ngure 3.21). Because of its nature, I c g ~ s in the same direction as the paths. Therefore, the generated electric fields will add, rather than subtract, as was shown for the differential-mode current in Figure 3.21 (Elc and E2c). Even a small common-mode current can produce the same level of radiated fields as a much larger value of differential-mode current. Based on the ~sumpfions previously stated and the treatment of differentialand common-mode currents outlined earlier, the incident electric fields in Equation (3.22) can be approximated as tE {I max = IE i •

m

a

x

( d i m t + tEi .

m

a x

(3.24)

(comm)t, •

where IEinax (diff)[ - 1.316 X I0-~4 !iD!f,__2(Hz) Ls d

(13.25)

IE i (comm) I - 1.257 X 10 ......6 / ~ f(Hz) L max d

(3.26)

and the units of the expression are volts/meter. Obtaining the valves of ID and I c can be pursued most easily by the measurement of I D with an oscilloscope at the frequency of interest and the measurement of Ic with a current probe, also at the frequency of interest. With the help of Equations (4.21) through (4.26), the total load voltage on the susceptible circuit of Figure 3.20 can be calculated using ~uations (3.18) and (3.27) as VL ( t o t a ! ) - (-~i)-Rf ~ _

( l + Ri)Rf V n l n R f

'

(3.27)

where Vn and I n have been defined in detail in Equations (3.21) through (3.26).

74

3.9

3. Attitude Control and Navigation

The Inertial Measurement Unit

The inertial measurement unit (IMU) is the he.a~ of a satellite's inertial guidance system. The objective of the IMU is to supply acceleration signals which are resolved into components of the desired coordinate system. The IMU is a platform containing two or three accetemmeters mounted on a system that keeps it stable in the desired coordinate reference frame. The other element of the. IMU is the gyro (or gyroscope): the sensor used to provide a stable reference axis in inertial systems. The basic gyro provides an electrical output signal propo~ional to the input angle or the integral of the input turning rate about the input axis, which causes a torque about the output axis 90" away, This section discusses two different types of gyros, as well as giving an introduction to the accelerometer.

3.9.1

MECHANICAL GYROSCOPE

The mechanical gyro is composed of a massive disk which is able to rotate freely about a rotational axis as shown in Figure 3.22. The disk is itself confined within a gimbals frame that is also able to rotate freely about one or two axes. ~ e r e f o r e ,

+

! I

Sp+nAxis [ _[_. ~ik ............. iiiiii~

......

•

........................

i

i

............... "

"

•

,I

,,,"'J~" r. . . . . . . . .

i

:+++;i

~

"-"+~"-I-.~1._

................

Input Axis App+iedTorque F i b r e 3.22

' ........ ----''--------'--'d

i

. J

~

~"'

,

i

Illustration of a mechanical gyroscope.

Output AXiS

Precession

3.9. The Inertial Measurement Unit

75

de~nding on the number of rotating axes, gyros have either one or two axes free. In gyros, typically (1) the spin axis of a tYee gyroscope will stay fixed with respect to space, provided that there are no external fomes to act upon it, and (2) a gyro can be made to deliver a torque which is proportional to the angular velocity about an axis perpendicular to the spin axis. When the wheel rotates, it tends to preserve its axial, position. If the platform of the gyro rotates around the input axis, the gyro then develops a torque around a pe~endicular axis, thus turning its spin axis. From Newton's third law of motion for rotating bodies, the time rate of change of angular momentum about an axis wilt be equal to the torque applied about the same given axis. tf a torque is applied about one of the: axes, and the angular speed w of the wheel is held constant, the angular momentum of the wheel may be changed only by rotating the projection of the spin axis with respect to the axis where the torque was applied. The applied torque is given by (3.28)

T = I w F~,

where [)~ is the angular velocity about the torque axis and I is the moment of inertia of the gyro wh~l. Good accuracy in mechanical gyros depends on eliminating effects that could cause friction, wheel imbalance, magnetic effects, and noise in the motors.

3.9.2

RING ~SER

GYROSCOPE

Gyros that have been built using lasers are based on the so-ca!led Sagnac effect as shown in Figure 3.23. As shown in the figure., two beams of light generated by a laser will propagate in opposite directions within an optical ring that has a refractive index of n; the ring has a radius R. One of the laser beam.s goes in the clc~.kwise direction, while the other beam follows in a counterclockwise direction. ~ e time light takes m travel within the ring is about t = 27rR/nc, where c is the speed of light. If the ring rotates at an angular velocity in the clockwise direction, the light will travel different paths in two directions. The clockwise light laser beam will travel a distance S = 2 ~rR + RFtt, while the counterc.lock~ wise beam will travel counterclockwise a distance S - 27rR ~ [~Rt. Hence., the difference between the paths is 4~,~R 2

AS . . . . . . . . .

nC

.

(3.29)

76

3. Attitude Control and Navigation

CCW

CW

2R

..........................................

~[

Figure 3.23 The Sagnac effect for building laser gyros.

In order to measure D., we must first d e t e ~ i n e AS. In a ring laser gyro (Figure 3.24), AS is measured by using the lasing properties of an optical cavity (which produces coherent light). For lasing to occur in such a closed cavity, there has to be an integral number of wavelengths within the complete ring. In order to compensate for the change in the perimeter due to rotation, the wavelength ,~ and frequency f of the light must change as

df = dA = ds. L

A

s

(.3.30)

If the ring laser rotates at a rate ~ then Equation (.3.29) shows that light waves stretch in one direction and compress in the opposite direction to meet the condition of an integral number of wavelengths about the ring, resulting in a net frequency difference between the light beams. If the two beams are mixed together, the resulting signal h ~ frequency

3.9. The Inertial Measurement Unit

Optical Coupler

Polarizer

Optical Coupler

CCW

77

Phase Modulator

~k[\, RingGYrO

Figure 3.24 A ring laser gyroscope.

4Aa f = Ans '

(3.31)

where A is the area enclosed by the ring.

3.9.3

BASIC FUNCTIONAL BLOCK D1AGRAM D E S C R I P T I O N OF A ~ S E R I N E R T I A L M E A S U R E M E N T UNIT

There are five major components of the laser IMU. The first component, of course, is the laser gyros themselves, which are located one per axis (gyro x, gyro y, and gyro z). The gyros supply angles and cu~ent data; the angle rates and torque data are preamplified, and with the use of comparators (which tracked the previous readings), the data is transmitted to a series of directional detection logics (in a gate array), where it is later processed. This sequence of events is described in Figure 3.25. The processor unit, which consists of a processor, RAM memory, EEPROM, and control logic, is in charge of processing the raw data from the gyros and transmitting it to the satellite's C&DH subsystem as shown in the figure. Also in the figure, we can obse~e the control logic sequence, which is multiplexed also using a field-programmable gate array (FPGA). An important pal1 of the IMU is the high-voltage power supplies (and associated control circuits) which are needed to drive the lasers within the gyro. The power supply ranges from 200 up to 2000 V; other power supplies generate the lower

78

3. Attitude Control and Navigation Gyro X

~Laser

A

Dry Control

iMU TxlRx Data Z ....:.

Control

T

)river Detector

iion

Comp, .......

,,.. . . . . . . . . . . . . . . . . . . ,,................................

.

A

Sig. Con&

GYRO Y

[_

--[---

..................................

I

'~ Rec

I

Processor

GYRO Z

~

Accumulators

Digitizer

Vel~ity to Force integrator

I

AJDCo,vl

Analog h~L~,,_i~ & Controls

Memo~ Access

. . . . . . . .

m

Figure 3.25

EEPROM

I/O Logic & Buffers

General 3-axis laser IMU bl~k diagram.

voltages (5, _+10, ± t 5 V) which are needed for the rest of the digimt and analog circuitry. Therefore, the an~lar rate (the useful output of laser gyros) is given by r=

ansi

4A"

(3.32)

In the ring laser gyro, a fiber beam splitter is formed such that there is very little coupling within the ring. When the incoming laser ~ a m is at the same resonant frequency as the fiber ring, the laser light couples into the fiber cavity and the intensity of the light being perceived by the detector drops.

3.9.4 ACCELEROMETERS The. function of the accelerometer is simply to sense linear physical accelerations and to provide a propo~ional electrical output signal. The linear accelerometer is of principal interest in inertial navigation.

3.9. The Inertial Measurement Unit

79

There are two basic physical principles involved in accelerometers. First, as Einstein theorized in his general theory of relativity, mass attraction is the same as true acceleration (i.e., dv/dt). Second, acceleration is referenced to an inertial reference frame. The accelerometer, therefore, is a sensor of a specific force--the resultant of gravitational force and inertial reaction force per unit mass. Specific force is dimensionally the same as acceleration, but of opposite sense. The basic workings of an accelerometer may be thought of as a damped, spring-restrained pendulum, as shown in Ngure 3.26. The pendulous axis (PA) is the arbitrary neutral position of the pendulum, and the input axis (IA) is the sensitive axis of the acceterometer and is o~hogonal to the output axis and the t~.ndutous axis. The axis about which the pendulum rotates is called the output axis (OA). The sense is such that OA rotated into IA by the right-hand m!e equals PA. One of the most common accelerometers is the pendulous gyro integrating accelerometer (PGIA). In the PGIA, a gyro with an unbalanced gimbals is used as an acceleration sensor. A servo-driven single-axis turntable (Figure 3.27) puts a mining rate into the gyroscopic accelerometer. Since the turntable rate is controlled by the gyroscopic acceteromemr output, this becomes a null-seeking device where the acceleration-sensitive toNue caused by the mass unbalance is exactly balanced by the gyroscopic toNue which results from the tumtab!e rate.

tA '~Ir.,.

Spring Gimbals • 7,., ..............L: .......................... ......._.,

t ~-o ...................................................................................... ~ "]

Damping

~ i !

~

[

Friction ..... ~OA .......................................... jr.

'l . . ,, !" 1 ' I ............................< > ] .................................. " V V ' ~ , ...................i.._.. . L N

PA

Figure 3.26

P~atform

Basic acceterometer description.

3. Attitude Conlrol and Navigation

Cmds,

/4

°, 1

!

~m~~~l...:.i.....!....Dampi .. ng

::i" .............................

...............

................ - .... ~*-;

......... M_O___!r_O___ ...........

"~,.L___J

'

Spring

"

;Friction

OA

.............................

........

~'

o.'om

............................................................................. l

Gyro Base Platform

Figure 3.27 Schematic diagram of PGtA.

Since the turntable rate is proportional to acceleration., the total turntable swept angle is precisely proportional to the time integral of acceleration. The output of each accelerometer is a current proportional to acceleration. The current is converted to digital pulses by the accelerometer's digitizer electronics. The pulses are then accumulated and read by the processor to determine the satellite's velocity and linear displacement. Figure 3.28 shows a block diagram of the accelerometer digitizer for one of the three identical accelerometer axes, The digitizer functions as a precision cun'ent-to-frequency converter. The operation of this converter is the foundation of the accelerometer.

3.9.5

NOISE ISSUES IN ANALOG.TO.DIGITAL CONVERTERS

At the heart of accelerometers and gyroscopes is the ability to receive and record sensor-acquired data and manipulate such data so. as to provide useful info~ation. In order to.. make that happen, analog-to-digital ( A ~ ) converters, such as those involved in current-to-frequency converters, must function properly and reliably

3.9. The Inertial Measurement Unit

81

with minimum noise interference. We now address the noise problems in A/D conve~ers in some depth. Noise in Aft) converters originates from two different sources: quantization error that is part of the data conversion process; and electrical noise, which includes the noise generated by the converter and noise that is part of the signal., as well as externally generated noise. Therefore, if the analog signal is later retrieved, it will represent not only the original recorded analog, but also the noise.. The total noise signal in the comparator of the A/D converter is the result. of the oDamp input noise, noise in the summing resistors, noise that is part of the original analog signal, and noise that is coupled, either from the power supply or from the external environment. As previously stated, the most basic noise is due to thermal resistance, also known as Johnson noise, given by the expression v,, = ~ ( f ) ,

(13,33)

where k T B R

= = = = V,~ -

1.381 × l0:23 J/K, known as the Boltzmann constant °C + 273.2 effective bandwidth in hertz resistance in ohms 0.129 (#V)

V/B(f)-R

Since the Johnson noise is just one of the listed noise sources, the overall noise from all other known independent noise sources can. be given by Vn(,,.,~a~ } = %/"V :~ I.......................... + V~':n2 .................................................................................... + " " • + V ~n'~ .....

(3.34)

~ e significance of the total noise in a high-resolution conversion depends on the magnitude of the quantum step (I least significant bit, LSB). Integrator

Sync .

............... I TorqueCurrent

! . . . . . . . . . . . . . . . .

i............../:..:.:~..4.%A~ ......._... _ ~

Comparator ~

'~7

Control

......................

Commands

Figure 3.28 Accelerometer digitizer electronics.

Putses

82

3. Attitude Control and Navigation

If the A/D converter is linear, LSB =

FSR 2n ,

(3.35)

where FSR is the full-scale range of the converter and n is the: resolution in bits. The equivalent input noise can be compared to this value in order to determine its digital significance: K = 2n V'~ FSR"

(3.36)

If the noise is Gaussian, the RMS value is that corresponding m the standard deviation or. We can relam, for example, to the case of the probability of a peak greater than 0.5, since K corresponds to a bit value of or. For example, the probability of a peak larger than 0.5 LSB will correspond m the probability of peak greater than 3 or, or 27%.

3.9.6

NOISE C O N ~ R N S IN. HIGH-SPEED SAMPLING OF ANALOG-TO-DIGITAL CONVERTERS

Let us look again in more detail at the internal noise of operational amplifiers; we will extend this to the case of high-speed analog-to-digital converters. ~ e r e are basically three noise sources in an operational amplifier: (1) a voltage noise which shows up differentially across the two inputs (inverting and noninverting), (2) a current noise in the inveaing input, and (3) a current noise in the noninverting input. Note that each of these noise sources is independent of the others. Figure 3.29 illustrates these noise sources. The voltage noise of different operational amplifiers may vary from 1 to

In.

Vn

in+ Figure 3.29

Noise sources at the op-amp input.

3.9. The Inertial Measurentent Illlit

83

20 nV/Hz. BJ'r op-atnps tenti to have lower voltagc noisc than JFET opamps. Voltage noise is normally specified on the data sheet. Currcnl noise usually varies much more widely, from 0 . 1 fA/Hz in JFET op-amps to s e v c ~ i lpicoamps per hertz in RJT. In voltage-fcedback op-amps, the current noise in the inverting and noninverting inpiit is uncorrelated (see Figure 3.29) and approxirnatcly equal in magilitude. In simplc BJT and JFET input stages. the noise cilrrcnl is the shot noise of the bias current, given by

where I, is the bias current in amps and 4 is the charge of an elcctron (1.6 X

10

''I

C).

Noise current is only important when it flows into a large impedance. hence generating a largc noise voltage. This means that the choice of a low-noise op-amp depends on the source impedance of the signal. At high impedances, current noise tends alwrtys to dominate. For example, the OP-27, which has a fairly high current noise (I pA/Hz): can best be used with low-impedance circuits. Figure 3.30 shows that different amplifiers arc best suited at different irnpedancc levels.

All horizontal scales HZ

All vertical scales

nv/\C-ii;:

10

100

1K

10K

Figure 3.30 Different amplifiers vs impedance levels. (Printed with permission from Analog Devices.)

3. Attitude Control and Navigation 3.9.6.1

Noise Figure in Op-Amps

The noise figure of an op-amp is the amount (dB) by which the noise of the amplifier is greater than the noise of an ideal (noise-free) amplifier under the same environmental conditions. It is a useful tool to know the voltage noise spectral density as well ~ the current noise spectral density. In operationa! amplifiers at low frequencies, the noise spectral density rises at 3 riB/octave, as shown in Figure 3.31. The frequency at which the noise starts to rise is known as the 1/f comer frequency and is a figure of merit. A currentfeedback op-amp may have as many as three l/f corner frequencies: one for its voltage noise, one for its inve~ing input-current noise, and one for its noninverting input-current noise. ~ e most efficient low-frequency low noise amplifiers have corner frequencies in the range. 1-10 Hz, whereas JFET op-amps have values in the region up to I ~ Hz. The RMS noise is obtained when. the noise spectral density curve is integrated over the BW of interest. In the. 1/f region, the RMS noise in. the bandwidth BW = fl ~ f2 iS given by

-dS) = K~/ln

(3.38)

f'

where K is the noise spectral density at 1 Hz.

NOISE

nV/qHz

White Noise I

""

..................................................................

t ....................................................................

ltf Figure 3.31 Frequency characteristic of op-amp noise.

Log f

3,9, The Inertial Measurement Unit 3.9.7

85

TOTAL NOISE OUTPUT

Consider the circuit in Figure 3,32, which is an op-amp containing three resistors (R s represents the source resistance). There are actually six noise sources: the Johnson noise of the three resistors, the op-amp voltage noise, and the current noise in each input of the op-amp. Each one of these elements provides a contribution to the: total noise of the amplifier output. In the circuit shown in the figure, C1 represents the source capacitance; it can also represent a stray capacitance or the input capacitance of the op-amp. C~ causes a breakpoint in the given noise margin; the capaci~nce C 2 is added to the circuit to obtain stability. C l and C2 cause the noise margin to be a function of frequency, and it ~ a k s at the higher frequencies. The DC noise gain is given by 1 + R2/R ~, whereas the AC noise gain is given by 1. + CJC1. The bandwidth is given by BW = 1/27rR2C2. The noise current of the noninverting input: In+ flows into R~ and gives rise to noise voltage ln+R s, which is DC and AC amplified. The op-amp noise voltage Vns and the junction noise of R s ( ~ ) are also amplified. The junction noise

C2

1!

Ii

..put

,,

v_

I

I:ls

............... r - , . Vn2

................................................

)

in+

Vns Figure 3.32 Modeling noise sources in a feedback amplifier,

86

3. Attitude Control and Navigation

of R~ is AC amplified over the bandwidth of l]27rR2C 2. The junction noise of R2 is buffered directly to the output with a bandwidth 1/2 ~rR2C2. Because of the negative feedback, the current noise of the inve~ing I n flows into R2, resulting in an amplifier output voltage of I n....R 2 over the same bandwidth of 1/2IrR2C 2. When we consider these six contributions, we can obse~e that if R s and R 2 are low, the effects of noise current and Johnson noise will be minimized, and the dominant noise will be the op-amp voltage noise. When the resistance is increased, both Johnson noise and the noise voltage produced by the noise cu~ent will rise. However, if the noise cu~ent is low, the Johnson noise will dominate the voltage noise~ Because Johnson noise only increases as the square, root of the resistance, while the noise current increases linearly with resistance, it is clear that as the resistance increases, the voltage due to noise current will become dominant. To calculate the total RMS output noise of the operational amplifier requires multiplying each of the six noise voltages by the appropriate gain and integrating over the frequency range of interest, as shown in Table 3.1.

Modeling Noise Sources in a Feedback AmpEfier

Table 3.1

No~e Source Expressed as a Voltage . . . . . . . . .

.

.

.

.

.

.

.

.

......................

::...

,

.............:..:

..............

Multiply by This Factor to Reflect to Output =

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

Integration Bandwidth .

.

.

.

.

Johnson Noise in Rs:

Noise Gain as a Function of Frequency

Closed Loop BW (Signal BW from Ground to Output)

Noninverting Input Current Noise Flowing in R~:

Noise Gain as a Function of Frequency

Closed Loop BW (Signal B W from Ground to Output)

Input Voltage Noise: "v'~

Noise Gain as a Function of Frequency

Closed Loop BW (Signal BW from A to Output)

Johnson Noise in R~~ ~V/'4kTRl

-R2/R ~ (Signal Gain from B to Output)

1/2'rrR2C2 (Signal BW from Input to Output)

Johnson Noise in R2:

1

lt2~R2C 2 (Signal BW from Input to Output)

Inve~ing Input Current Noise Flowing in R2: In-R2

I

1/2~R2C 2 (Signal BW from Input to Output)

3.9. The Inertial Measurement Unit

87

The noise gain for a typical second-order system is shown in Figure 3.33. Notice that the large noise voltage will. be determined by the high-fiequency paix where the noise gain is C~/C2~ The noise which is due to the inventing input current noise, R l, and R 2 is only integrated over the bandwidth 1/2 rrR 2C2. Figure 3.34 shows the noise model for a typical second-order system. In high-s~ed op-amp applications such as in A ~ converters, a first-order system simplification is suitable; the noise gain plot is usually flat up to the close-loop bandwidth frequency. This means that all noise sources can now be integrated over the closed-loop op-amp bandwidth as shown in Figure 3.34. In high-s~ed current feedback loops, the input noise voltage and the inverting input noise current are the dominant contributors to the output no"~se, ~ as shown: in. Figure 3.35.

3.9.8

PROPER POWER SUPPLY DECOI~SOLING .IN OP-AMPS

If the supply of an op-amp changes, its output should not really change, but in reality it does. The term power supply rej~tion ratio (PSRR) is defined as

........................

-----~...

GAIN dB

• "

SIGNAL= \ \ \ . \ INPU

R2~ R 21^'^~

..............

I I

f2 Figure 3.33

SIGNAL \\ GAIN

NDWIDTH

[ 1 I

fcL

CL BANDWIDTH

"\,\,

"" Loaf

Noise gain inverting signal gain for second-order system. (Printed with permission from Analog Devices.)

88

3. Attitude Control and Navigation

1

R~

................................................. )I .........................................................

"t__

I ~

l

"'

t "--.

I ~ ......................."-:e.

I

V ~"-~,~.~......... [. ...........~,~N ....... 11 + ~

[

J

............................

Jn.'~'

'..

/

~ "" , C 2 ..N~

I. ....................................................... !........................... ,>~

. . . . .

2 R2~1 v,srf 2 ) + t .+ PL + C'~/2~2 ( ~ ' c s ~ .

) + t.? R~

Where fu = Op Amp Unity Gain Bandwidth Frequency fcl = Closed Loop B a ~ t h

.................

NEGLECTED

'.

VoN.

Figure 3.34

.O,S~aA,N

7"1

"

1

=~ , 1+C 2

Noise model for a typical second-order system, neglecting resistor noise. (Printed with permission from Analog Devices.)

Rf Vn

[ [

..................................................................^ - , / X / X .

.........................

+v

/

•

In-,

Figure 3,35

Noise model in high-speed current feedback scenario,

Vo

3.9. The Inertial Measurement Unit

89

follows: If a change of A volts in the supply produces the same output change as a differential input change of B volts, then the PSRR on that supply is A/N. It is because the= PSRR of op-amps is frequency dependent that op-amp power supplies must be well decoupled. For op-amps at low frequencies, we can use 1 ~ 5 0 #F capacitors for each supply, provided, that no more than 10 cm separates the capacitor from the op-amps. At high frequencies, each IC must have every supply dezoupled by a low-inductance 0..1.-#F capacitor with short leads. The capacitors must also provide a return path for high-frequency currents in the op-amp load. An example of proper low- and high-fi'equency use of decoupling capacitors is shown in Figure 3.36. 3.9.8.1

Bypassing and Grounding to Avoid EMI Noise

As is well known, there is no real ground in operational amplifiers, but rather a "virtual ground." In digital-to-analog conveners which provide current to an op-amp, the current does not really return to ground, but instead returns to one of the power supplies. In order to reduce the impedance in the high-frequency

C3

<10cm

c ..............................................................

C1 ~

',/¢ L E A D L E N G T H

,, . . . .

MINIMUM

..L = L A R G E - A R E A v GROUND PLANE

....

, C2: o,........=~,,,',,',,,,,, ......

<1:0era ~I -Vs

Figure 3 . ~

'=

L O C A L I Z E D HF DECOUPLING, LOW INDUCTANCE C E R A M I C , 0.1 p.F

C3, C4: S H A R E D LF ELECTROLYTIC,

10- 50~F

"

Low- and high-frequency decoupling for op-amps. (Printed with permission from Analog Devices.)

90

3. Attitude Control and Navigation

current path, a bypass capacitor should be connected to allow He cu~ent to return from both power terminals to the ground provided for the digi~l-to-analog converter. If the digital-to-analog converter h ~ active grounding problems, its own power supply input will also need to be bypassed. In the use of bypass capacitors, care must be exercised as to their location. In Figure 3.37, a bypass capacitor is being used in a very ineffective manner. Here the op-amp drives a load impedance Z L. However, the ground remm is quite long to the power-supply terminal. Also, the bypass decoupling capacitor returns to the power supply through another long path. The result is that the return path for the load current is longer than the supply lines powering the op-amp. Not only is the bypass capacitor ineffective; it may also contribute to noise problems. Figure 3.38, on the other hand, shows a more efficient scenario, in which the decoupling capacitor connects using the shortest path between the load return and the load voltage control element. In this figure the decoupling capacitor connects to the _ V pin of the op-amp to the low side of the load, providing the most direct return path for high-frequency currents and bypassing them around ground and power buses. In a large system it is often not practical to depend on a single cornrnon point for all analog signals. When this is the situation, it is necessary to use a differential amplifier to translate and differentiate between the two ground systems. A signal is translated, from one ground system into a simplified signal which is referred

ZL

v'~/'-.,------

Grounded Load

Power Supply Ground DC/DC CONV. V

Figure 3,37 Ineffective use of de~oupling capacitors.

3.9. The Inertial M~surement Unit

L

+

.oo

-- -

Cdecouplingpling

.............................. i:............~.................................. =

:.................. ~; ---- ~:=-V

,,~ .................... ,......

91

~

,....................................................... ~----

-~-

DC/DC CONV. V

~ .--~,

F i b r e 3.38 Effective use of decoupIing capacitors.

to a different ground signal, as shown in Figure 3.39. The common-mode rejection of the amplifier and a resistance ratio match are used to eliminate the effects of voltage differences that exist between the two grounds, It is important to power up the op-amp from the power that is available at the load side of the circuit and/or to decouple it with respect to the common terminal load. As is well known, an op-amp converts a differential input signal to a singleoended output signal,

Load Circuit Power Should Atso Be Used

DAC DCOM

i

ACOM

................. l

.......

,.,,,,,

.:

:

.

.

.

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......... R° ........................ .

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Rf .

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.

Figure 3,39 The use of a differential amplifier to eliminate the effects of commonmode voltage.

92

3. Attitude Control and Navigation

In most op-amps, the differential-to-single-ended conversion is done with respect to V..... and the resulting signal drives an integrator. The integrator is used to frequently compensate the amplifier, and at V_. It can be seen that the integrator acts as a unity gain follower for fast signals applied to its noninverting input. The result is that signals which are applied to the V~ terminal have their high° frequency components conveyed directly to the output. Fu~hermore, signals having frequency components above the amplifier's closed-loop b~dwidth will be transmitted from V~_ to the output with very little: alteration. As shown in Figure 3A0, if the op-amp which is used as a subtractor is bypassed to the same common line as the input signal, high-frequency signals associated with that common line will show up as part of the output signal. If the ground noise includes high-frequency noise (.as in digital circuits), the common-mode, rejection will be useless. However, if the op-amp supply te~inals are referenced to the output-signal common, no extema! signals will be coupled into the integrator. Any ground noise a.p~ars as a common-mode input signal and is diminished by the common-mode rejection of the amplifier. Often, since noise rejection performance of the subtractor depends on the matched source and f~dback resistance ratios, it can only be used in specific situations, and not in those situations where the source impedance cannot be controlled or the source impedance is really high.. When. such is the case, ground noise can be avoided by the use of an instrumentation amplifier as shown in. Figure 3.41. 3.9.9

M O R E F U N D A M E N T A L S A B O U T OP-AMP GROUNDING

As we have shown, since two ground points are seldom of the same potential, the difference in such ground potential Vg will couple into the circuit as shown in Figure 3A2. In the figure, Z~I and Zt2 represent the impedance of the two traces connecting the signal Vs to the op-amp, and Rg represents the ground impedance. The input voltage Vi~ is equal to V~ = v~ (neglecting all other intrinsic noise sources). To eliminate Vg, one of the ground connections should be eliminated, most simply, the ground connection to V~. Assuming that the input impedance of the op-amp Zin > > Zt~' + Zt2, and ZI~ is the load impedance, the noise voltage in Figure 3.42 is given by

It is often necessary" to protect high-gain amplifiers from strong e!ectromagnetic fields by enclosing them inside a metallic shield. In Ngure 3.43, stray capacitance

3.9. The Inertial Measurement Unit

93

Ri V+

,

Input Signal

Output

(a) V~.

Ri

LOADS RF

Ground Noise

Cdec,

vA,/,,,. Ri V+

input

Signal

Output

(b) LOADS C:dec.

~.c,,£j. Figure 3.40 (a) Improper and (b) proper decoupling of subtractors using op-amps,

provides a feedback path from output to input, and if this feedback is not eliminated, the op-amp will most likely oscillate. The shield connection that will eliminate unwanted feedback paths in the operational amplifier is one where the shield is connected to the amplifier common terminal.

94

3. Attitude Control and Navigation

w,f ........................

,,,

"......................................................................... +

•

"

Cdec.

s . . E : : N : S E ............ ! N

Signal

•

UT

.......................................................................................

ADC -

Ground Noise

.........

"

T

"k.

[ACOM ....DCgM_::

...............[ "

Cdec}

Figure 3.41 Instrumentation amplifier sep~ates ground systems.

Vout Z t2

Vin

. / /

Figure 3.42 Com~monimpedance coupling in op-amp grounding.

When the connecting wires to the inverting and noninverting inputs are shielded as shown in Figure 3.44, and if the signals are low frequency (A > > L, where L is the size of connecting wires from the signal source to op-amp), the shield should be grounded at any one point when. the signal circuit has a single-point ground. If that single-point ground is at. the amplifier common, then the shield should also be grounded at the same c o - - o n , as shown in the figure. In the figure, Vg~ represents the potential of the amplifier common terminal above earth ground, and Vg2 represents the difference in ground potential between the two ground points.

3.9. The Inertial Measurement Unit ;:~ .::...

: - ...............

:

o_,

,_.._., ..,.. ..:....

.=, :

.... ========================................... === ---:

95

..........................

~,~.

-E -~ C1

C2 \ Vou t

I

C3 .....................................................

T

Figure 3.43 Guard shields of op-amp,

'----_./7"

....1

.

]

\7 Figure 3.44 Cable shield one-end grounding of op-amp circuit.

If the single point ground is at: the signal source, then the shield should be grounded at the same signal, ground, as shown in Figure 3.45. When the connecting wires to the inverting and noninverdng input are shielded and if the signals are high frequency (3, < < L, where L is the size of the

96

3. Attitude Control and Navigation

%

c~

Vout

%2

Figure 3.45 Cable shield one-end grounding at source end of op-amp.

connecting wires from the signal source to the op-amp), then the shield should be grounded at both ends, as shown in Figure 3.46. If a shield is used against electromagnetic interference, the proper grounding is shown in Figure 3.47.

c1

%

Figure 3.46 Cable shield grounding at both ends.

%

7

Figure 3.47 Maximum shielding of op-amp against EMI.

%ut

Chapter 4

4.0

Satellite Power Subsystems and Noise in Power Electronics

Introduction

The high reliability needed in space systems, especially for missions of long duration (more than 5 years), has spurred the development, of lightweight but reliable power systems. In this chapter we address the various elements of satellite power systems, including energy sources, energy converters, energy storage, power condition and distribution, and certain control aspects. Since the emphasis is on noise concerns in such power systems, considerable space will be dedicated to the analysis of how noise issues are addressed in spacecraft power systems.

4.1

Solar Energy and Power

The amount of electrical power required on board a satellite is dictated by the goals or mission objectives, such as the o~rational requirements of the payloMs, the antenna and telecommunications requirements, the dam rate, and the satellite orbit. In communications satellites, the power requirement range from 5 ~ t.o 2000 W, depending on the channel capacity. A satellite power system includes the following components: 1. A primary source, of energy, such as direct solar radiation, nuclear power generators, or chemical batteries 2. An. assembly for converting prima~ energy into electrical energy 3. An assembly for storing the electrical energy to meet peak energy demands or eclipse conditions 4. A system for conditioning, charging, discharging, regulating, and distributing the generated electrical energy at the specified voltage levels The basic configuration of a satellite power system based on a solar energy source is shown in Figure 4.1.

97

98

4. Satellite Power Subsystems and Noise in Power Electronics

Battery Charge Controller

Loads

Battery Charger BatteryCharger Fused PWR to Loa~

Battery

Power Electronics Assy y Gimbals

Figure 4.1

4.2

Basic configuration of a satellite power subsystem.

Solar Cells and Radiation

Sol~ cells furnish most of the long-term power supply for satellites and other spacecraft. The radiative energy output from the sun derives from a nucle~ fusion reaction in which, each second, about 6 × I0 l~ kg of H 2 is converted to He., with a net mass loss of 4~0 × 10 3 kg. By the Einstein relation E = mc 2, this is equivalent to about 4.0 × 102o J. This energy is radia~d mainly as electromagnetic radiation in the 0.2 to 3 ~ m wavelength range. The intensity of the solar radiation in free space at the distance from the sun to. the earth is defined as the solar constant with a value of 1353 W/m 2, A typical solar ceil works as a simple p - n junction with a single bandgap energy of Eg. When the cell is exposed to the solar spectrum, a photon with energy less than/~, makes no contribution to the cell output. However, if a photon has energy over Eg, it contributes an energy Eg to the cell output and the excess over Lg is wasted as heat. Consider the energy-band diagram of a p - n junction under solar radiation shown in Figure 4.2. The solar cell is assumed to have an ideal I-V characteristic. The equivalent circuit is shown in Figure 4.2 also. The source I~ results from the excitation of excess carriers by solar radiation; I a is the diode saturation current; and R L is the load resistance. Ec and Ev are the conduction and valence bands, respectively.

4.2. Solar Cells and Radiation

99

hv

/ E v ..................................... J

~

hv

....

,.

,dl .....

R,L

Figure 4.2 Electrical parameters of a solar cell. The I - V characteristic of such. a device is given by I = I d (e q V / x T -

1 ) - I s.

(4.1.)

From this equation, we obtain the open, circuit (I = 0) voltage: ~ c = qKT In (~'Is+ I . ) ~ K---T ( / ~In ).q

(4.2).

A plot of this equation is given in Figure 4.3, which is the current-voltage characteristic of a solar cell under illumination. For a given I s, the open-circuit voltage increase logarithmically with decreasing saturation current l d. The output power is given by P = I V = IdV(e q v / K T -

I)

-

IL~

(4.3)

The condition for maximum power can be. obtained when d P / d V = 0, or

(

Imax - I L 1 1

')

fl%ax.

Vm~,~,x= V ~ . - ~ ln(1 + flVm.ax),

(4.4a) (4.4b)

100

4. Satellite Power Subsystems and Noise in Power Electronics

Ii i(mA)

V

- Volts

0

+ Volts

Figure 4.3 Current-voltage characteristics of solar cell under illumination,

where fl ~ q/KZ The maximum power output Pmax is then given by 1

Pma×=q Voc-~In( 1 + flVmax) - ~]1 s.

(4.5)

The early designs of solar cells resemble that of Fignare 4.4, which lasted until the 1970s. It was then realized that sintered aluminum along the rear improved performance, probably by a combination of guttering and. the fo~ation of.heavily doped near interface known as a back surface field. Further developments included finer, more closely spaced contact fingers and the use of better antirefle~tion coating. Pyramids on the surface, ~ shown later in Figure 4.5, reduce reflection so that, after antireflection cooling, the cell looks like black velvet. ~rami.ds also couple light obliquely into such black, ceils, allowing abso~tion closer to the surface.

4.2.1 SOLAR CELLS IN SPACE SYSTEMS Solar arrays designed for LEO and GEO orbits will probably require low to moderate power. Low mass and a long life will also be necessary.. ~ e low mass is directly related to the size of the spacecraft itself; the lighter the solar arrays,

4.2. Solar Cells and Radiation

Top Metal Finger

\

101

Antireflection Coating N-Type

P-type

Metal Contact

Figure 4.4 Sol~ cell design on silicon.

the more cost-effective the spacecraft will be, because it will be lighter and will require less propellant for correction maneuvers during its lifetime. A~ay lifetimes are determined by the rates at which the solar array electrical outputs degenerate with time to the point at which the satellite can no longer fhnction properly. Though there could be mechanical, contamination, and thermal re~ons for solar a~ay degradation, the most significant cause of electrical degradation is the constant bombardment of the ceils by natural charged panicles in the radiation environment. The extent of the damage depends on the material of which the cells are made (i.e., silicon or gallium arsenide) and the ability of the cells to anneal as the damage continues. As previously stated, the orbit location (GEO, LEO, MEO) affects the power system design, which includes not only the solar array, but also the energy storage batteries and power distribution systems. Battery charges and discharge cycles differ greatly, depending on the sunlight-to-eclipse ratio of the orbit. The battery storage subsystem must be designed to provide all. the satellites with the needed. power requirements during eclipse and to fully recharge during the sunlight portion. "When the satellite's inclination is low, as in low earth orbits, the sunlightto-eclipse ratio is about 2 to I: this means that for every hour of sunlight., there is approximately 30 minutes in eclipse. The life of a battery is greatly affect~ by the depth of discharge and the rate of charge. The battery usefulness is mainly affected by the depth of discharge and the rate at which that discharge occurs. Battery usefulness is measured in terms of watt-hoursNg. It is important to realize

4. Satellite Power Subsystems and No~e in Power Electronics

102

inverted Pyramids

Light

!

Metal

P+

Oxide

High Resistive Silicon

Figure. 4.5 PyramiNsha~d silicon oxide for attracting sunlight.

that the depth of discharge and the rate of charge directly d e t e ~ i n e the extended life of a battery. A smaller battery, for example, can provide reduced mass for a satellite, it will also have a depth of discharge that is much larger (and thus more dangerous) than that of a larger batte~,~, Geosynchronous satellites make only 365 orbits per year around the earth (compared to 6000 orbits for LEO satellites in 1 year) and undergo only 90 eclipses with a maximum duration of only 1 hour. Therefore, in GEO satellites the depth of discharge is smaller than in LEO satellites. Figure 4.5 shows some features in modem solar cell design, such. as the use of "inverted pyramids" in silicon oxide to attract the sunlight.

4.3

Solar Arrays

Solar arrays (see Figure 4.6) for satellites come in flexible deployable arrays made up of hinged panels. The hinged panels are made of a sandwich, of two face sheets bonded to an aluminum honeycomb core, after an insulating sheet is

4.4. The Space Environment and Radiation Damage to Solar Cells

panel hinges solarpanel , / " ~ sE,arce,,s mast ................ ' I I

...................

L 'l..... ,

I

- ~ ' I

i

I

(

/

103

t

SolarArrayI Rotations I

......

~

.................................................................................................................... ~.......... ~............ ! .............t ...............

i

m

.................

i

i SPACECRAFT \ \ %

Figure 4.6 An example of a solar array panel.

bonded to the panel surface on which the solar cells are to be placed to provide an electrical insulation between the cells and the panel. Solar array size is an important issue for small LEO satellites. Increasing the size of a solar array to satisfy power requirements is often not a popular option, since the increased moments of inertia on orbit restrict the maneuvering capabilities of the spacecraft (it. may even affect the programmatic solar array rotations to detect the sun). The most practical option for increasing array power is to use advanced, higher efficiency solar cells, such as gallium arsenide. However, the cost of using advanced solar cells compared with the cost of silicon cells is an important factor that. must be weighed against the cost of making changes in the satellite design that would consume less power.

4.4

The Space Environment and Radiation Damage to Solar Cells

One of the most important factors in solar cell degradation is constant bombardment of solar cells by charged particles generated by the sun, Figure 4.7 shows the region, of space in the general vicinity of the earth known as the magnetosphere. The radiation environment near the earth is composed mainly of electrons and protons trapped in the geomagnetic field, along with other temporao, types of radiation (atoms, ion.s) associated with solar flares. The effect of galactic cosmic rays is minor. The earth is responsible for the radiation belts near the earth due to the earth's dipole magnetic field. The radiation belt traps and holds the chm-ged

104

4. Satellite Power Subsystems and Noise in Power Electronics

Energetic Electrons and Protons /\

\Magnetic Lines

\

'i

\

6,000M

•',

i

i''~'\.,,

1/

\ l. I

1 i

I/

\ I

\"\ ""\',,\

/

t \

/"

THE VAN ALLEN BELTS

Figure 4.7 Space environment for solar cells. pa,aicles for long periods of time, forming a low-density plasma in the inhomogeneous field. The plasma is dynamic in nature, with a constant flux of particles. Solar flares are the source of particles entering the pl~ma. Most of the solar flares' protons and high-energy electrons pass through the plasma; lower-energy plasma particles are trapped in the earth's dipole field~ In Figure 4.8 we see the distribution of charged particle types in near-earth space. The densities and energies of the trapped protons m ~ e them the predominant cause of solar ceil degradation below four times the radius of the earth. The inner-zone electrons have less of a contribution. In geosynchronous satellites, the effect of trapped protons is very small and the main damage is caused in reality by trapped outer-zone electrons. The energies of these particles trapped range from a few MeVto hundreds of MeV in solar flares. The particles resulting from sol~ flares (solar flares are usually in 11-year cycles) cause about the same degradation of solar cells due to trapped energetic particles as those degradations that occur slowly over the same period of time. Trapped electrons result in a steady, commutative loss of power, where~ the solar flare protons can cause a large fluence of energetic panicles and charged particles.

4.5. Switching Power Supplies and Converters

Charged Particle Regions

l

2.8 3.8 ~ t 1 I Trapped I I LPro.tgns l........................... 1 il ~' InnerZone

1

5.0 1

105

Solar Flare Protons

t

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

/ ' t I I

2

3

4

5

6

7

8

9

10

11

12

Earth Radii (Re) Figure 4.8 Distribution of charged particles near the earth.

4.5

Switching Power Supplies and Converters

Future satellites will become more modular in nature: instead of the satetlite's main power subsystem providing the regulated power to all the loads in the spacecraft, each payload will accept unregulated power from the spacecraft, power subsystem, and then use its own switching-mode power supply m regulate the power and convert the initial spacecraft voltage to lower voltages as needed. In this section we will address some of the fundamentals of switching-mode power supplies and the intrinsic noise sources generated by them which could affect other payloads in a spacecraft. We will also address several noise issues related to this kind of power supply. The switching-mode power supply (SMPS) is a class of power supply which uses electronic switching m process electrical power. Switching operations require very little waste of energy, and when the switching is done at high frequencies, the size of transformers, filtering elements, and feedback circuits can be minimized considerably. Switching-mode power supplies are very common today and will eventually replace most linear-power-supply applications, except when conducted interference needs to be minimized.

4.5.1

INTRODUCTION TO POWER CONVERTERS

A DC-DC converter receives a DC input and produces a controlled DC output. The three basic types of DC-DC converters are the buck converter (which, with

106

4. Satellite Power Subsystems and Noise in Power Electronics

an isolation transformer, is called a forward converter), the back-boost converter (which, with an isolation transformer, is called a flyback converter), and the boost converter. Variants of the forward converters are the push-pull converter, half-bridge converter, and full-bridge converter. The buck converter is shown in Figure 4.9. The ele~tronic MOSFET switch is driven on/off at a high frequency (5-500 kHz). The duty cycle of the FET switch controls the DC output voltage Vo,t. The output filtering capacitor CL is used to smooth out the ripple component of the output voltage due to the highfrequency switching. The energy storage device is the inductance, and if it is chosen large enough, the current 1L through it will be "smoothed out," while the pulsating current Ii wili be of square or trapezoidal shape. In order to regulate the output voltage, a control feedback loop is added m the main converter. An error amplifier and a pulse-width modulator (PWM) are the main ingredients of the feedback loop. Assuming that Vout = AVe, we have

If

Aft

Vou., = A(Vref -- fl "l/out)

(4.6)

A V°ut - Vref 1 + Aft"

(4.7)

> > 1, the output voltage will be approximately equal m Vout = l/tel//3-

Duty Cycle 7 D J

PWM

i

*

i

CONVERTER_._I

IV,: I

! Vc

!

(4.8)

........l~rror .................... ]=........~===A............................................................... 1

141----Ampl

VO -

i Ve

(A)

l .......................

I

± c . ,

i

1

J Figure 4.9 Buck converter illustration.

..............]

1

R

VOU,.

4.5. Switching Power Supplies and Converters

107

Vot,~ is then really independent of the DC input voltage ~ and the toad current. The above SMPS is an example of a voltage-mode controlled l~gulator. We make the following assumptions: 1. The buck converter is in a steady state. 2. ton = DE where D is the duty cycle of the PWM and to,~ is the time the

FET switch is turned on within a switching period (T). 3. The inductance L is large, enough that the inductor cun'ent I L will not go to zero during the time the FET switch is turned off (continuous-mode operation). 4. Q is large enough that, within a switching cycle, the change in ~{,ut is v e u small. We can now show the waveforms of i c, i i, &. VD, and ~,ut for the buck conve~er (Figure 4.10). The buck-boost converter with voltage-mode controlled feedback is shown in Figure 4. t i. The voltage and currant waveforms in continuous mode of operation are as shown in Figure 4. I2. The boost-convertor circuit is shown in Figure 4.13 with its voltage-mode controlled feedback circuit. The voltage and current waveforms in continuous mode of operation am as shown in Figure 4.14. The Cuk converter is a cascade of a boost converter followed: by a buck converter. Figure 4.t5 shows a Cuk converter, and also shows how a Cuk convermr may be decomposed into a boost converter and a buck converter. The two converters represented in Figure 4.15 share the same electronic switch and flywheel diode Dr. The storage capacitor is used as the output filmr capacitor for the boost converter; it can also be used as a battery, providing power to the buck converter. The load resistance R L shown in the boost converter represents the loading effect of the buck converter. The main advantage of the Cuk converter is that its design lends itself to providing a ve U smooth input current IL2. The Cuk conve~er combines the merits of a boost converter, with its smooth input current,. and a buck converter, with its smooth output cu~ent. However, in order for the Cuk conve~er to achieve this advantage, both the boost and buck parts must operate in continuous mode (i.e., the current is maintained in Df during the entire period of D T < t < T when the. switch is turned oft). Cuk converters usually produce less noise or ripple noise because they have slower rise and fall times. A f o ~ a r d converter is a buck converter with an isolation transformer. Converters coupled to transformers are used in space systems in the following situations: I. Power supplies with multiple outputs. An isolation, t r a n s f o ~ e r with multiple output winding can eliminate the need for multiple converters.

108

4. Satellite Power Subsystems and Noise in Power Electronics

.................... Vt

DT

T

V

3DT

DT

3DT

2T

t

t

.....................................................................................................................

DT

v°u~ L

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

T

3DT

t

Vout = DVi

-y

DT

T

t

Figure 4.1.0 Voltage and cu~ent waveforms of the buck converter,

2. Some. converters are required to provide step-up or step-down. Transformers can provide that kind of solution. Using high-frequency transfo~ers has some adverse effects on the highfrequency switching w a v e f o ~ s : 1. The transformer inductance results in an inductive shunting current. 2. The leakage inductance in transformers may produce large current spikes during switching periods, especially at high. frequencies.

4.6. Noise in Switching-Mode Power Supplies

Duty Cycle = D •

1

.

[

PWM

r-:~

,o

[ :....................... ........... __ __ -

~'~

Amp,. L ] 1t3

..........................['

~11Ve

[~

_h T vref

2,

......................

~N~ERT~

1

t....... I"

E~ror

" - -........... . . . .

,II

L

......................

i ................................... ,_.,J°

109

. . . . .

,,......,. T ............... ,)

voo,

J

Figure 4.11 The buck-boost converter,

3. Transformers introduce additional loses in the power supplies such as eddy currents, copper loss, and hysteresis loss. An example of a forward converter is shown in Figure 4.16. The transformer T2 magnetically couples the output of the PWM to the power MOSFET transistor. The transformer T 1 provides the functions of voltage step-down and galvanic isolation between the regulator outputs and the Vm input. The operation of the forward converter is shown in Figure 4.17. A buck-boost converter with an isolation transformer is ~ o w n as a flyback convener and is shown in Figure 4.18. Figure. 4.19 shows the idealized current and voltage waveforms. R e transfo~er coupled push-pull converter is a push-pull version of the forward converter. The push-pull converter is shown in Figure 4.20. The idealized w a v e f o ~ s for the circuits axe shown in Figure 4.2i.

4.6

Noise in Switching-Mode Power Supplies

Though this has not yet been discussed, the input voltage m any power converter ~n is that which is taken after the EMI filters. In this section we first address the noise of the input of the power converters and how it can be filmred out; we then address the noise source generated within the converter.

110

4. Satellite Power Subsystems and Noise in Power Electronics

IL

I"

dlL/dt = V~ 1 L

t

dIc fdt = -Vo / L

A1 L =

]

................................... L ~

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

DT

(1-D)T

DT

(I-D)T

- . = ~ ...........................

:

...............................

_ :======..:,,_o

................

._ = . . . . . . . . . . . . . . ~ . l ~ .

t

I

|

k

= = . . . ... . .. . ... ... . .. . ... ... . .. . ... ... . .. . ... . . . . . . . . . . . .[ . .. . . . . . . . .

DT

t

.

.

.

.

.

.

.

.

.

.

.

.

.

~ - . - :

(I-D)T

.....

~ m,,,-

t

Vo = D V~ / (I~D)

voo( DT

T

t

Figure 4.12 Voltage and cu~ent waveforms for the buck-boost converter.

4.6,1

INPUT NOISE

The basic requirements to the input stage of a power converter are as follows" 1. The maximum ripple noise voltage must ~ within the allowable range acceptable to the DC-to-DC convener. 2. The input inrush cu~ent, which is the input cu~ent when the DC main bus from the spacecraft is fed in to the assembly, must be limited to a reasonable value so that it will not damage the circuit.

4.6. Noise in Switching.Mode Power Supplies Duty Cycle = D

i

12~

................................................................

_ + . - _......... "~ ~ P ~ 4 = ..~. ~ ' ....

IL

.-~ L

.........................................

,d~

.................................. ........] ............... +

'

..........................

E,+,+o,-

Vref

+j_+I,L :

..................

C~N~,"~T~ ~

+'4+

#, .......,........ :.~.!~+~, ~,

.......... , .....

1"

++A...................+--~)

~AmLnl('

111

.

,

+

Out

+,

J

Figure 4.13 The boost converter.. 3+ The conducted interference that gets into the DC main spacecraft bus should be minimized to avoid conducted interference in power buses. 4.6.2

INPUT RIPPLE NOISE

A tow input ripple voltage must be maintained in a converter; otherwise, the mgulator will not function properly. A regulator can only tolerate a limited input ripple voltage. There is always a limit on the ripple in the DC input voltage of the converter. In order to maintain a low ripple voltage them is a. need for a large input filtering capacitance (7, as shown in Figure. 4.22. A common practice is to allow the ripple to be 30% of the peak value of the. minimum allowed DC voltage on the S/C power bus. For example, if the average power bus voltage is 28 V and the minimum allowed voltage (or undervoltage) is 22 V, then the maximum ripple voltage allowed into the power converter should not be more than 0+30 (22 V) = 6+6 V, which translates to about 35 V maximum, allowed voltage. Figure 4.22 shows the input circuit to a typical switching power supply. The minimum capacitance C r.equimd is given by 1P(.

1 + )+

(.4.9)

where P is the output power of filtenng or input power to the switching+mode power supply (SMPS), f the DC input ripple fl'equency, and V,nin the minimum bus-allowed DC voltage.

112

4. Satellite Power Subsystems and Noise in Power Electronics

I,.. FET off

li

...,,................................. ,,,DT.. ,..T ton I toff

,

3~T

....

DT

T

......

I S0T

dIL / d t = V~ / L

[

li=lL

2T

| dlL/ dt = (Vi- Vo) / L

3DT

t

t

...................................................................................................................................................................................................

DT

3DT

Vdt DT

T

Vout

2T

3DT

IIIPV"

t

t

Vo =V~/(~-D)

DT

T

t

Figure 4.14 "Voltageand current waveforms for the boost converter,

4.6.3

INPUT INRUSH CURRENT LIMIT

As soon as the FET switches, closes, or is turned on, the instantaneous charge current flowing into the c~acitor C can be, quite large because of the large voltage difference between the DC input voltage and the capacitor voltage. This

4.6. Noise in Switching.Mode Power Supplies

Duty Cycle- D_ .

~

113

A::::::: Ve ,

.

i

Vref

h

L2

+

Df

(a).

RL Vout

t

L.

. . . . .

t

~l',tV~~R

_!l~.~._.r',ccra_ ................... 2,

Vo

._j

'

RL

v,:,ot.

(c)

(b) Figure 4.15 The Cuk conve~er.

may damage the SMPS input. To prevent possible damage, a triac in parallel with a current-limiting resistor R1. is inserted into the input circuit as shown in Figure 4.22. The current-limiting resistance R1. limits the maximum inrush cun'ent to a reasonable value. When the capacitor C has be~n charged to its maximum capacity, a continuous gate drive: is applied to the triac to turn it on. to shunt the current in R~. 4.6.4

CONDUCTED I N T E R F E R E N C E

Switching-mode power supplies produce internally conducted interference. The conducted interference is transmitted through other power buses into the spacecraft, and it must be. blocked before it affects the inputs of other switching-mode power supplies which are exacting clean power. The coupled inductors LI and L2 and the capacitors Cx and Cy are shown in Figure 4.22 and used as a line filter to stop conducted interference from polluting the DC main bus of the spacecraft. The. polarity of coupling between L~ and L2 is such that the magnetic effects of the input DC cu~ents cancel, each other out. The inductors act as a. common-mode choke, for common-mode r~ection. Since

114

4. Satellite Power Subsystems and Noise in Power Electronics

I'lv,i:3i~ D,T~~I 0. R'L

T1~I-

_I~. ,,................... r l

PWM .

.

.

.

.

.

.

.

Error I Ampl. Ve _L. " ....... 13 [~ "r Vref .

. . . . . . . . . . . . . . . . . . . . . . . . . . . .

Fiffure 4.16 A forward converter.

Ll and L 2 are wound in such a way that they have significant leakage inductance, the leakage inductance can also reject differential-mode current. Typical values of L 1, L2, Cx, and Cy are Li = L 2 = 2 t o 5 0 m H Cx = 0.i to I / z F Cy = 2200 p F t o 0.033/xF. The actual values depend on the range of frequencies the filters are supposed to suppress.

4.6.5

SOURCES OF INTERFERENCE GENERATED IN THE SMPS

During their n o d a l operation, power semiconductors generate high-frequency disturbances because of their high repetition rates, which range up to several kiloheaz. The spectra of power semiconductors reach up to several megaheaz.

4.6. Noise in Switching-Mode Power Supplies

Vgs

FET ON

1

FET OFF

!. . . . . . . . . . . . .lOT .................. .osI .......... ..........................I.,

......................

.........

A Vp T vp = ~o_

22 T

'in ~~n=Vin ns/np[

.

.

.

.

.

.

vp =-~n

t

,...................................T

.....

!

t ¸

..

T~

DT VD - ~ ns/np

I ..............]

Vout~

.

T

DT

,•

t

....

DT

.... r o ~

VD

115

T ...............

t

I

_

1

Vout = D ~fn ns/np

t Figure 4.17

4.6.5.1

Voltage and current waveforms for the forward convener,

EMI from Diodes

A diode is really a switching element, since it acts as a short circuit with forward bias and an open circuit with reversed bias. The switch on operation of a diode is shown in Figure 4.23. The transition time from off to on is given by to. The

116

4. Satellite Power Subsystems and Noise in Power Electronics

li

D,

[V,n ... ................ ¸

.................................................... + 'd L

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

Vout

T

r vos .

+

.

J ............P ................................. ]~c ............

WM

I E r r 0 r .......'.................................. .. ,

Ampl

ve

Figure 4.18 The flyback converter.

forward current I d g ~ s up quickly, A relatively high on-state voltage also appem:s on. the diode. This voltage falls back again to a nominal value in the time interval tf. The voltage spike is really a broadband, emission. During the switch-off operation, the broadband emissions are actually much higher. The switch-off voltage and cu~ent are shown in Figure 4 . ~ . The onstate current decreases to ~ r o at. to; the current becomes negative because of the stored charge in the f o ~ of minority current c ~ e r s in the depletion region. The reverse or negative, current will continue until a time ts while charged carriers are still present. The current reverses quickly to zero in a time tr when ca~iers are depleted. Since: the reverse cu~ent can be quite large and. the reverse time v e ~ small (I #sec), high voltage transients with wide broadband frequency spectra can appear in the inductance of conductors and connecting circuits, R e emissions of diode rectifiers at switch-off can. be reduced by using RC snubbers connected in parallel with the diode rectifiers. 4.6.5.2

Noise from Silicon.Controlled Rectifiers

Like diodes, silicon-controlled rectifiers generate high-frequency noise during both switch-on and switch-off operations. In silicon-controlled rectifiers, the noise levels are higher during switch-on than at switch-off. The voltage and current curves of silicon controlled rectifiers are shown in Figure 4.25.

4.6. Noise in Switching-Mode Power Supplies

Vds l

FETON -- FETOFFt ...............

= . . . . . . . .

........

..................................................................................

Vp I vp =Vin v~

DT

[

---=-=-

.. ~-=-~

~.................

T

I ......

T+DT

f

Vp=-Yout np/ns

...........

F ~

i

~

,,

'°t

DT

.21

T

-

.........OX

I - - - - 1

I ........................... i ..............................

"T

..............................................................

!

,

T+DT

T+DT

-

Y

.... vo I. ........................................................

t

1..........................

.............................. [ .....................................-t ~.....................................!.............................

t.

V°ut t

117

D,

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

t ...........

i...:: ........... "

"

"

"

.............................................

_.................

,--.~-----._,.

--~,., ;~------~ ...........

;~ ....

III1~.

y

t Figure 4.19

Voltage and current waveforms for the flyback converter.

np

T

,,j.........

.............

i .......................................................................................... I ..... ~ yi~_r

h ...................................................................... t

np Figure 4.20

The transformer coupled push~-pulI converter.

V•o u t

C

•

i RL

11.8

4. S a t e l l i t e P o w e r S u b s y s t e m s

Vgsl A

1

........... t! ............................... t2

Vgs2 ~ - - ~ T -

Vds2

el

Electronics

DT

I

Vds I

a n d N o i s e in. P o w e r

l

_

t3

.

.

.

........

.

t

................ 2 ..................... 2 ~'il;iil-;ii;ilZi2111;i;i;i " l ..............................................

=2~n

iii ~-~ '-ii

" i~iiiii i .......................................................... ~

t

&

t

T

....=:_2%~....

l

i.......~.......

................................................................................. l

I.........

=Vin~.n_slnp

A

t

t

T__ .........vm nsfnD ............................ ":.!~D~

Vou~

I...................i ..................!.................... ...........I.................... .....

........................

t

_

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

=, 2P.....~:n....nsln.p ...............................................................................................................................................

t Figure

4.21

"Voltage a n d c u r r e n t w a v e f o r m s f o r the p u s h - p u l l converter,.

119

4.6. Noise in Switching-Mode Power Supplies FET Switch

.......... j

P

_..........

•

1

....................................................................................................................................

_,.-.. i S M P S

Ouput

! u

..................

Figure 4.22 Typical input: filter in most power converters.

The signal a~ives on the control electrode at time. t,,. The depletion region becomes active after a given delay time t a, After that, the voltage in the siliconcontrolled rectifier collapses very quickly. This collapse of the anode~athode voltage is responsible for generating the broadband high-frequency disturbances.

Vd

to

i tf

t

I I

Id

i

I I I

1

t ..................................j

.......................................................................................................................

Figure 4.23 Switched ON operation of a diode.

120

4. Satellite Power Subsystems and Noise in Power Electronics

ts

tr

t y

i

Figure 4.24 Switched OFF operation of a diode.

4.6.5.3

Noise From Power Trans~tors

From the point of view of output noise, the interference generated by power transistors is similar to that of silicone control rectifiers. Figure 4.26a shows the typical voltage and current waveforms of switch-on of power transistors. Figure 4.26b shows the curve of the collector-emitter voltage and collector current of a power tr~sistor at swimh-off. The switch-off time begins at to; then the collector current continues to flow until a given time ts, called storage time. In this interval the charged carriers will be removed from the depletion region. After t~, the collector current Ic falls to. zero. The fall time of the collector current is very sho~ (10 nsec to 1 ~ ~e~), and this explains why the noise specmJm of power transistors is very wide. Because of the repetitive nature, of the switching pulses in diode rectifiers, silicon-controlled rectifiers, and power transistors, the EMI noise generated by power electronics is that of a pulse train. The

4.6. Noise in Switching-Mode Power Supplies

..................

121

::::::::::::::::::::::::::::::::::::::::::::

t ....................

_ , ,

,

,

:,

,,

...........

- ,

.........................................

I

--

..,:---

,

......

.......... , , : : , , , , ,

,, . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

,

td

e

.............................................................

t t' Figure 4.25

Silicon-controlled rectifier noise.

electromagnetic emissions from semiconductor circuits are broadband and coher~ ent. For studying EMI noise from semiconductor circuits, the amplitude spectrum function is the most useful tool. ~ t us consider the constant switching of the power FET transistor, the diode rectifier, or the silicon-controlled rectifier in a switching-mode power supply. The switching waveform will be that shown in Figure 4.27. ~ e magnitude of this spectrum can be calculated from the Fourier t r a n s f o ~ to produce IVnl = spectrum components =

sin(neff~ T) (n ~'r/T)

sin(n rrt r / T) (n rrt r / T)

where n ¢ 0 is t,he~"h a ~ o n i c number, T is the wriod of the periodic waveform,

4. Satellite Power Subsystems and Noise in Power Electronics

122

IB Switch is ON

t

(a)

i

I

to

IC

td

ton

F i b r e 4.26 "Voltageand current waveforrns from a typical ~ansistor at turn-on (a) and turn-off (b).

and we have assumed that tr = tf. ~ e spectral bound for this trapezoidal w a v e f o ~ is given by Figure 4.28.

4.6.6

INTERFE~RENCE PATHS IN SWITCIHNG.MODE POWER SUPPLIES

Now that we have identified the sources of interference in SMPS and managed to express such noise in a quantitative manner as shown in Figure 428, we attempt to describe how the noise is propagated within these switching-mode power supplies. Consider the f o ~ a x d c o n v e ~ r shown in Figure 4.29. The conducted e ~ s s i o n s (or conducted noise) in switching-mode power supplies is a mixture of both differential-mode and common-mode noise. These two modes come from different sources and axe dealt differently in the fine filter. The

4.6. Noise in Sv,itching-Mode Power Supplies

tg A

123

Switch-OFF

.~ .--:.............---::...... ........

=.,,.,,.:, ..

,.,,..,~_,::

..

:,.

(b)

&

Vc~

& IC

I

~c

= •

I

to

f

=

~r'-~_

ts

F i b r e 4.26

i, V(t)

VCE . i

I

.........................

toff continued

..........,,............../-.........

%

tr

tf

Figure 4.27

T

Switching wavefonn for serrdconductor circuit.

t

124

4. Satellite Power Subsystems and Noise in Power Electronics

Spectral Power dBgWm

. . . . . . . . . . . . .

0 MHz

f- - I c m !

\

D

~

v,°. + c , l l-..L

Chl

......

,i,I | . ).~, ~ r( IN' ],_.[___ ,

! , .....

.

.

.

.

.

.

.

.

.

.

Spectrum of trapezoidal waveform.

/"

. . . . . . . . . .

.

500 MHz

frequency (MHz)

Figure 4.28

.

.....!..... H I I '

~

/

'

""

'

L

I=,I '

-~ oft eLI ! )

........................................................................

I

,cm Figure 4.29

--~ Forward, converter with parasitic elements.

4.6. Noise in Switching-Mode Power Supplies

125

differential-mode current Idm in Figure 4.29 is the normal c.u~ent provided by ~n and follows the intended forward and return paths. The differential-mode currents are equal in magnitude and opposite in direction; they are the functional or desired cun'ents on the line. The common-mode currents (l,=m) are undesired currents. They are not needed for the functional performance of the switchingmode power supply. These common-mode currents are the result of parasitic couplings (C hl, Ch2. Cq in Figure 4.29) between elements of the switching-mode power supply and ground paths. The result is that common-mode currents most often, follow unintended paths, adding in some cases to produce cu~ents of higher magnitude than differential-mode currents. These higher cu~ent magnitudes are capable of radiating, producing not: only high-magnitude electronic fields due to "antenna' '-type radiation configurations, but also significant conducted current, which is seen as conducted noise and affects the proper operation of switchingmode power supNies. The total current in an SMPS is then the sum of these two currents, as shown in Figure 4.30. Notice, however, that at difl%rent frequency ranges, mostly only one. of the currents is dominant. This is an important issue because tailoring the design of an EMI filter m cancel both common-mode and differential-mode currents must be accomplished one component at a time. 4.6.7

A STUDY IN THE PROPER DESIGN OF GROUNDING FOR SMPS CONVERTERS IN PCB

Conveners that are used in switching-mode power supplies are usually well designed internally m maximize efficiency and minimize output noise. However,

Idm is dominant

...................................................................................................................................................................................................................... I.I ..~ ..

f(Hz)

Figure 4.30 Differential and common-mode current magnitudes in SMPS.

126

4. Satellite Power Subsystems and Noise in Power Electronics

deficiencies in the distribution of power and grounding within the PCB where the SMPS converters will be located can negate many of the advantages provided by these converters. Inductive and capacitive effects on the way SMPS converters are used in a PCB are one of the major causes of high conductive emission in PCBs. Furthermore, changes in grounding layouts can affect these inductive and capacitive, effects. Finally, CEM tool.s can be used in the modeling of these parasitic effects. ~ e emphasis in this section is on design principles. 4.6.7.1

Introduction

The switching~mode power supply is a class of power supply that makes use of electronic switching to process electrical:, power,. Because ideal switches do not dissipate power, the SMPS can be designed to have a high. efficiency. In the SMPS a high switching frequency is used, and the size of the transformer and filtering circuits can be minimized. Because of these great advantages, the SMPS has become the power processing unit of choice in low-power circuits or in circuits where interference must be kept to a minimum.. The heart, of an SMPS is a DC-m-DC converter. The converter accepts a DC input and produces a controlled DC output. The three basic types are the buck converter, the boost converter, and the buck boost, converter. For each of these converters there is. an electronic switch that is driven orutoff a high frequency (5-..5..00 kHz). It is the duty cycle of the electronic switch which controls the DC output voltage V,,ut. An output filtering capacitor Cout is used to smooth out the tipple components of the output voltage resulting from the high-frequency switching. By adding a feedback circuit in a converter, the output voltage of the converter can be regulated. In each converter circuit there is an energy-storage inductance L which can be chosen large enough so that the current in it is substantially smoothed. Because SMPS DC-DC converters are v e ~ sensitive to input noise, such noise must be filtered out as much as possible. For that [email protected], an EMI filter is chosen as shown in Figure 4.31. The figure shows a single EMI filter (FM-461) at the inputs of several DC-DC converters which are used to supply different voltage levels for different loads. The filter EMI modules are especially designed to reduce the input line reflected tipple current of DC-DC converters. These filters are intended for use in applications of high-frequency swimhing (I 00 kHz). ~ e filters are capable of reducing the input tipple current by as much as 40 dB within the frequency band of 1.00 kHz to 1 ~ MHz. Inside the DC-DC converter, good design mchniques are applied m minimize the output noise from such conve~ers. It is well understood by designers that a

4.6. Noise in Switching-Mode Power Supplies FM-461 .

.

.

.

DC CONVERTERS

.

.

iii ~n

Vout

.

.

.

.

•

Dvin

Case GN !II~N CaN

127

-

. . . . .

1

Vout

RL

ii) Case GND OUT C l ~ IN CMN

OUT CM

Multiple Units Allowed Up to Rated Output Current

RL

..........,7 ..~..::.::. .~__ i,i ?~

v,o_ ...........1................................ t

i .........!.............................. .. t.....

T lN5649A

T

T

i ..... ~

,

Case GND

~ Output ............................................................................-E: ..................................... ... lnput

Common

Figure 4.31

FM-461 Schematic (Simplified)

Common

EMI filter serving a cascaded line. of DC-DC converters.

clean output voltage is essential for the proper functioning of ICs, and especially analog devices and circuits, which are highly susceptible to buswoltage noise~ Therefore, in DC-DC converters it is as important to control input noise as output noise. We first outline some principles of limiting input noise, and then concentrate on our main subject: limiting output noise from PCBs where conveners are ibund. 4.6.7.2

Transient Effec~ in SMPSs

In the simplified circuit of a switching-mode power supply, an en'or amplifier compares output voltage V.,,,twith a reference Vr,,fand controls the duty cycle, D, via a pulse width modulator as shown in Figure 4.32. The output capacitor Cou, is represented by its equivalent circuit that includes the equivalent series resistance (ESR) and the equivalent series inductance (ESL). When we have a load step 8I, current through the choke inductance L cannot be instantly changed. There will always be a finite time t n e e d ~ for L to accommodate aI, given by the expression LaI t > (EnDmax) _ ~,,~ _ i/diode,

(4.10)

where Dm. ~ is the maximum duty cycle and Vd~oao is the diode's voltage drop, The choke current Ichoke slews to the new load current, but before it does that

128

4. Satellite Power Subsystems and Noise in Power Electronics

F

lchoke

t,

I Lwir~rac.~ Flwire/trac_.e out I

t L ~-~

i ................

,.,

i

/

Jr

E

1

t

L.

t

,A,

. . . . . .1.....:.... . . . . . . . . . . . . . . . . . . . . .L,:<J:_: . . . . . . . . . . . . . . ............ . . .i. . . . . . . . . . . .

' I

O,

|

o

47

e e

! ErrorAmp~

Vre f

SMPS

I

PCB Loaas

Figure 4.32 Simplified diagram of an SMPS with output capacitor model (ESR & ESL).

/load flOWS through Cout. This results in an output voltage deviation 8Vout that may be as much as 8Vout < ESL(dl!oad) dt + ESR &

(4.I I)

-

where dlioJdt is the load's current slew rate (A/sec).The SMPS's Co.t acts as a reservoir for these current transients. The delay that is observed is compounded by the wiring and possible long traces in the PCB. As can be observed in Figure 4.32, traces have self-inductances and resistances, and when /load changes from Faraday's lave, Lwircma,=c will cause an initial voltage deviation 6V given by

(4.12)

[Jaez = ~ Lw-irelt--rac¢s(dll°ad) dt "

Furthermore, Rwireltracewill cause an input voltage drop as/toad slews. The time that is needed to change a current through load wires/traces in a PCB is given by t = td~lay + tri~,

(4.13)

where taeiay is the SMPS delay time and trisc is the time needed for lwireltr,ces to catch up to the load current, given by

"-'~

= tdeiayl'/v

/(

L\

dl l . o . < ~l d t 1 m~ _,,o~d ) -~wir ~

/ --

dt J

(4.14)

4.6. Noise in Switching-Mode Power Supplies

129

where Vm~x is the maximum output voltage during the transient recovery of the supply. The output load will experience a dip of as much as (4.1.5) •

.

\.

CMad

/

A computer simulation of a circuit of the type shown in Figure 4.32 using SPICE can show the effects of load wire/traces, inductances, and external capacitance, as shown in Figure 4.33. 4.6.7.3

Proper Grounding to Suppress ~ a n s i e n t Effects

Since parasitic inductance and resistive effects in the loads of SMPS are greatly responsible for the transient effects as shown in Figure 4.33 (the Fourier transform of Figure 4.33 wilt show a series of discrete frequencies of conducted emissions), efforts to minimize such transient effects will go far toward reducing the conducted and radiated emissions that are so common in power-supply buses.

1

69 O

__L_L_ t

_ '_L_L_L_q_

I

!

I

Ill

O~ C t-"O O tlJ

,

,

I

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w

I _ L _ L _ i i

L _ L _ L _ L _

_ L _ L _ I I

I t

10.0

t~

L _ L _ L _ L _ I

O

,,

I

I

I

i

I

l

.............I ..........................!,~.................................. ! ..............................l

Time (pS)

!

!

90.0

Figure 4.33 Load transients resulting t~om modeling output loads of an SMPS.

130

4. Satellite Power Subsystems and Noise in Power Electronics

Designers of PCBs that are m accommodate SMPS conveners must control the noise emanating from the on+board converter so that it does not interfere with other Systems circuit~ or propagate into the main power bus+ Converters are usually designed to pass CE and FCC radiated and conducted emissions. However, this is really not enough, since emissions at the PCB level must also be contained. Conducted emissions containment is seldom provided within the latest high+power modules. One+ reason is that leaving it out gives the designer greater flexfbility in meeting design requirements; it also reduces cost and realestate.+ requirements. Therefore, good board layout is essential for minimizing the amount of noise an on+board converter conducts or radiates. Good board layout is essential for maximizing power efficiency from: on-board converters to other PCB loads. Ideally, the board should provide wide power paths routed closely together in parallel. In addition, close-loop areas:, which can+behave as antennas, should be kept to a rrfinimum. To help shield other circuitry from the radiated noise in a fast-switching power train, board designers should avoid running signal lines under the converter. Common-mode noise, which is coupled through the capacitance between components such as heat sinks and transformer isolation windings, ap~ar between frame ground and the converter's input conductors. Differential noise appears across the input conductors.. Common-mode noise showing high frequency content can be routed back to the on-board con.ve~er by ceramic capacitors placed between input and output conductors and the case ground. Lower-frequency differential mode noise can be diminished using ceramic capacitors placed close to the converter between • e input leads. All of these bypass capacitors should be placed as close as possible to Me converter to ~nimize loop ~e.as. Ngure 4.34 shows the proper placement, of capacitors in DC-DC converters. Once bypass capacitance has been placed in a smart configuration to reduce common-mode noise from the DC-DC conveners and PCB, we look at the ground and power layouts from the CAD system to see what improvements would further reduce the conducted and radiated emissions. In Figure 4.35 we see the layout of power and ground planes corresponding to.+the DC-DC converter and PCB schematics shown for Ngure 4.34. N.gure 4+36 presents experimental data for conducted emissions obtained for a PCB based on the design of Figure 4+35 (other components of the PCB are not shown). There are three basic rules commonly used by designers for containing the noise generated by the power module: (1) Return the noise current to. the source using as short as possible a return loop, (2) re/Juce the impedance of these loops

4.6. Noise in Switching-Mode Power Supplies

Input 0,01#F

GND Plane +Vin +Vin

O.01uF

GND Plane +Vout

SMPS

Corn "Voot

GND Plane

131

28 #F

Load

- 28 ~F

Load

- ........................... 0

GND Piane.

O

O O O

----i O.01~F

................................+;7 .. °

%

O'0tUFT , SMPS.. ~jom j ......... +Vin

0~0!~F ~ne

~~

___T

.............. T

__

o

TO.O~F

Load

Figure 4.M Proper placement of bypass capacitors in DC-DC convegers.

by reducing inductance and increasing capacitance, and (3) identify alternate routes and suppress them by adding impedance. 4.6.7.4

Using Parasitic Extraction Tools for Optimizing Power and Ground Layout for DC-DC Converters in a PCB

The flow cha~ in Figure 4.37 shows a brief outline of the procedures followed for designing PCB using high-level hardware description languages such as Verilog and VHDL. Notice that an integral part of this modeling process (e.g., for designing a PCB with DC-DC conveners) is the use of parasitic extraction tools to perform a complex parasitic extraction process and obtain parasitic effects such as inductances, resistances, and capacitances in the PCB. This procedure is used for obtaining power-plane and ground-plane parasidcs. As the flow c h ~

132

4. Satellite, Power Subsystems and Noise in. Power Electronics

+Vout +Vin

(28v) -Vin

~ = CeramicCaps. = Electroi~ic

Caps.

-Vout

-Vot~

OutputGround +Vout

-Vo~ Figure 4.35

Ground and power plane layouts for DC-DC conveaer,

4.6. Noise in Switching-Mode Power Supplies t40

133

28V L E A D i..........ITHilii~

130

............

t20

•

. .

-

...............

/

tt0 100

~

~

........ ............a~

<"

..........i,,~i

80

60 50

E

<

~...........-b,

70 ar ~, ~t ~abieC£ N ~~~-~IC0-C~2!099-01 Fig~re 29 Crew CempaM:ment

~

:2ZZ

40 30 2O

!.................. ............

-t0

10

-20

~' i a n d w

~3o L _ t0 t

t0 2

JPL E M C Lab

Figure 4,36

)0

L

......

10 3

i

~iooo t00

104 t 05` Frequency (Hz)

~

~ O C ©

106

! 07

108

Data Acq. Vet. Z0

Conductedemissions for a PCB with three DC-DC conveners with ground and power layout as shown in Figure 4,35,

shows, final timing simulations am performed to estimate performance. The procedure can be repeated several times for each change of the power- and groun&ptane layout until an optimum design is obtained. A newer and more optimized layout than the one shown in Figure 4.35 is illustrated in Figure 4.38 after the optimization procedure of Figure 4.37. Finally, some conducted emission measurements were made on the new design in Figure 4.38. Figure 4.39 shows some improvement at a couple of frequencies, where the emissions were reduced to the. noise, level. Further optimization can. be obtained if the same techniques am applied at other parts of the: PCB beyond the DC-DC converters. Transient effects and common-mode noise current in SMPS are directly respon.sible for the conducted emissions often seen in the 30-kHz to 100-MHz region. This section has shown some of the origins of these transient effects and modeling associated with such effects, It has also been. shown that minimizing p~asitics in the power and ground planes of DC-DC converters will not only diminish possible transient effects, but will diminish common-mode noise as shown in conducted emissions.

134

4. Satelfite Power Subsystems and Noise in Power EleetrorAes Physical Flow

PCB Design

Flow

Functional Simulation

Modeling Flow VHDL/Verilog (for Digital)

'~-"-.,, Specified

Analog Models

,.

Synthesis

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

Timing Analysis

Analog Physical -" Models for Parasitic Effects /

Gate Level Simulation

VHDLNerilog ~r

Anatog Behavioral Models

Place-and-Route

SPICE-like Tools

Final PCB Models

Fibre 4.37 Flow chart for PCB design.

4.7 Energy Storage: Batteries In any spacecraft power system that uses solar radiation, the batteries within the spacecraft are the. most important, source of continuous power.. Depending upon spacecraft orbital location and ~lipses, batteries provide the needed power for all spacecraft operations. The batteries are charged dunng the sunlit portion of the orbit and disch~ge during the eclipse.. In the case of spacecraft in LEO, the nurnber of eclipses increases as the attitude decreases. For example, for a 5508 km orbit, there will be approximately 15 eclipses per day, or about 5500 per year. When the spacecraft comes out of eclipse, the power output of the solar a~ay is much higher than the. steady-state power output. The extra power can then be used if the battery is capable of being charged at. high rates.

4,7. Energy Storage: Batteries

135 +Vout

+Vin

(28v) -Vin

-Vout

Ceramic = Caps. 0 = Electrol~ic Caps. -Vout

~Vout

Figure 4.38 Optimized design of DC~DC conve~er power and ground, planes.

A storage cell is made when two unlike metals are dipped into an electrolyte material, as shown in Figure 4.40. A celt is an electrochemical device that stores energy in chemical form; this chemical energy is then converted to electrical energy during the discharge. The magnitude of the electrical energy inside batteries depends on the cell voltage (or potential), the kind of electrochemical reactions, and the size of the cells. The proper selection of storage ceils depends on several factors: spacecraft orbital altitude, orbit inclination, eclipse power requirements, operating temperature,, length of mission, and peak. power requirements. For spacecraft in LEO at a typical 550-kin orbit altitude, there are approximately 15 eclipses per day with a maximum shadow duration of 36 minutes for

136

4. Satellite Power Subsystems and Noise in Power Electronics I40

28V LEAD

t 30 t20, 1t0 100 ~-

:;:3

~3 "~ Q

E <

80 70 60 50 40 3O 20

-lo -20 -30 101 JPL EMC Lab

t 02

10 3

10 4

t 05

Frequency (Hz) "

"

~0 6

t 07

10 8

o ~ Acq.Ver~2,O P~o~ter Vet, 2,0

Figure 4.39 Conducted emissions for a PCB with three DC-DC converters having the ground and power layout shown in Figure &37.

every orbital period of 96 minutes. The charging and discharging cycles average about 5500 per year. When the spacecraft comes out of eclipse, the solar array has a cool temperature and therefore produces more power. As the solar array gets hot, it produces less electrical power. The storage cell must have the capability of delivering power at high rates and be able to sustain a large number of charge--discharge cycles. The cells must have a high recharge efficiency and good seals to prevent the loss of electrolyte. One of the most commonly used batte~ cells at present is the NiH2 battery. This cell consists of a catalytic gas negative electrode coupled with a nickel positive electrode. Electrochemically, the reaction at the positive electrode is coupled with a reaction of the negative electrode. At the negative electrode, hydrogen is displaced from water by oxidation reaction of the negative electrode. The most improved packaging technique used in these cells is the common pressure vessel (CPV). A common pressure vessel cell encloses a stack of ele~trodes connected in series; each stack is parallel-connected inside. Figure 4.4i shows the CPV arrangement for a NiH 2 battery. Figure 4.42 shows the arrangement of such a battery within the spacecraft. A battery consists of series-connected storage cells of the required ampere-hour

4.7. Energy Storage: Batteries

1137

Electron flow

_L anions (-)

Negative

electrode

Electrolyte

Positive electrode

0

cations (+)

+

Battery Charger

_

Figure 4.40 Electrochemical description of battery electrotyte.

capacity. To yield a particular voltage, the battery is charged, either at a constant rote or with a varying cmlent. When the battery reaches its end-of-charge, it is then normally trickle-charged. The charge rotes and operating temperatures are related to each other and are selected as a function of such factors as charge-discharge cycle life, temperature control system limitations, and the time required for replenishing the battery from a given depth of discharge. 4.7.1

BATTERY CHARGING AND POWER CONTROL

Battery charge control (including end-of-cha.rge and depth-ofodischarge control) is obtained, by time integration of the. battery" charge and discharge cun'ents (ampere-hours), computation of the battery efficiency, computation of the battery depth of discharge, and the measurement of battery voltage when compared

138

4. Satellite Power Subsystems and Noise in Power Electronics

Positive I~ Terminal __ __

•-.~Thermal

Flange

Seal

..... Electrode Stack

.....................h......Assemby ........................................

Negative

Terminal

Figure 4.41 Common pressure, vessel battery cell in NiH2 battery. against computed temperature-compensated voltage limits. Monitoring circuits for undervoltage and overvoltage protection are also pan of proper voltage charging of the battery. In Figure 4.42, an electrical power regulation b l ~ k diagram is shown. The figure illustrates the solar array currents from the spacecraft with their respective partiM shunt regulation. The p ~ i a l shunt assemblies are activated to propo~ionally shunt excess solar array current to maintain the regulated spacecraft bus voltage. The figure Mso shows the battery power control electronics, consisting of a boost voltage regulator (BVR). Notice that in most spacecraft power subsystems considerable redundancy is added to protect against inadve~nt failures. Finally, the B VR is attached to the fuse board assembly which protects the power bus with redundant fuses.

4.7. Energy Storage: Batteries

["

]["

ilk

')

ibu s

7[

Vj

'l t t I

Ishunt

' ( i

t !

1 1 |

vBus

i

[

iRshunt

~ <

Rshunt

139

>

<>='R.shunt <

9

ba~e~

i

i

I l I

Figure 4.42 Battery-solar array arrangement: in a spacecraft.

The power control electronics transfers power from the solar array and batteries to the spacecraft and payload. The battery charge assembly charges the batte~ during periods of batte~ use. If solar array current is insufficient, a BVR is activated m supply power from the batteries, The BVR is propo~ionally reduced as solar an'ay power is available to provide the required bus power, then turned off. For solar array power in excess of load requirements, excess current is used m charge the batteries. For solar array power in excess of load and battery requirements, the partial shunt assemblies are activated to propo~ionally shunt excess current. Battery charge regulators provide charging rates as selected to provide safe operating conditions.

4.Z2

INTRODUCTION TO WORST.CASE ANALYSIS OF POWER ELECTRONICS

In spacecraft electronics for power circuits, components are subject to large stresses over a period of several years. Many of these stresses awe caused by wide temperature ranges (sometimes - 2 0 to 90°C) as the spacecraft goes from sunlit to eclipse conditions, several times a day. Power electronics is subjected to these large temperature variations because it is here that some of the most significant power dissipation occurs.

140

4. SateUite Power Subsystems and N o i ~ in Power Electronics

Life cycles in spacecraft electronics also contribute to quality and requirement variations of spacecraft power electronics. As months and years go by, component parameter variations become more wonounced+ Resistors, for example, start experiencing resistance variations. Aging of parts will also become a factor in ~ f t variations in component values+ The combined effects of environmental and component parameter variations, which are tied to the length of satellite missions, and other aging effects cause considerable stress to all spacecraft electronics, but more especially to power electronics+ Such stresses in power electronic circuits produce the same, and in many cases, greater, adverse consequences than any noise or interference could+

Chapter 5

5.0

Command and Data Handling Subsystem

Introduction

The command and data handling subsystem (C&DH) provides the communications to and. from the spacecraft. The C&DH is the "brain" or central processing unit(s) of any spacecraft. It is where commands are received from the ground, processed, and executed to perform a variety of functions within the spacecraft. It is also where commands are internally generated for executing specific or redefined maneuvers. ~ e C&DH also behaves as an information center, receiving raw data from the different sensors (e.g., attitude control, thermal, propulsion) and measurements from payload instruments. The data is processed (compression, formatting, and storage) and then modulated for final transmission to earth as telemetry signals. The C&DH subsystem also stores all flight, software (which generates the internal commands) and fault protection software which is executed in several portions when the spacecraft experiences certain faults and are detected. In this section we will address the fundamentals of a C&DH subsystem and the fundamentals of command/data transmission. The senti.on will end with interference.

5.1

Brief Overview of Satellite Command Systems

In very simple terms the satellite command and data handling subsystem can be described as that of Figure 5. I. The base band signal which carries the encoded command message up to the spacecraft is modulated with a carrier RF signal. (Career signals are usually S-band (1.6-2.2 GHz), C-band (5.%6.5 GHz), or Ku-band (14.0-14.5 GHz).) A receiver and modulator in the spacecraft captures the signal, demodulates the command message, and then delivers the message to the decoder, usually in the form of an encoded subcarrier signal in the frequency range 14-.2.0 kHz. The decoder decodes the original command message, and it also provides a lock or enable output, together with a clock or timing signal which is synchronized to the message stream. The command logic checks the command message for fidelity. Several check=in routines are employed m validate each command. Command validation is critical in the design, of a spacecraft 141

142

5. Command and Data Handling Subsystem Cmds TLM •

C&DH

AACS

~

Subsyste ~

Cmds, Data,

and Interrupts

L__ Interface Bus

.

Fuse Pwr TLM ~,& Ant. Pos. & Ant, Cntrt.

Power

Subsystem

Mechanism .

Burn Wires .

.

.

.

Uptink Cmds S&A D a t a TLM

Telecomm, Subsystem

Cmds..

Figure 5.1 Brief overview of command and data handling subsystem.

command system: invalid commands such as those corrupted by noise may have serious consequences for spacecraft subsystems. When the command has been validated, the command logic drives the appropriate interface circuitry so that the command is executed. The command decoder examines the subcarrier signal, and the message is then detecte& The command decoder also provides a clock to the command logic. The clock tells the command logic when a bit is valid on the serial output. The clock shifts the command message into the command decoder and also provides a lock signal. The lock signal tends to indicate to the command logic that a command is going to "be shifted out of the decoder on the serial data line.

5.2

More Derailed View of C&DH

The basic requirements of the C&DH subsystem in a spacecraft are as follows: 1, Provide compu~tional capability for spacecraft command and control 2. Provide communication, interfaces for uplinked commands 3. Accept housekeeping telemetry from the spacecraft subsystems and payload 4. Format science data and telemetry data into transfer frames

5.2. More Detailed View of C&DH

143

5. Provide data storage which will allow simultaneous recording and playback 6. Provide gimbal-dfive el.e~tronics for driving redundant motors for solara~ay and antenna deployments 7. Provide independent firing of burn wires for releasing solar arrays and high-gain antennas 8. Provide control of main engine and reaction engines for spacecraft co~Tective maneuvers 9, Provide independent firing of pyrotechnic values 10. Provide electronics to interfaces in the AACS subsystem 11. Provide power switching for thermal control 12. Generate power conversion, and regulation to several, interface buses. A more detailed block diagram of a command and data handling subsystem (C&DH) is shown in Figure 5.2. Following is a brief description of each of these assemblies, 5.2,1

FLIGHT PROCESSOR COMPUTER

The elements of a flight processor computer are shown in Figure 5.3. The microprocessor is part of the flight, processor board; other boards included are Redundant Uttrai ~ S t a bOsci l eBator

I C&DH I Unitlnterface

....................... Solar Array Dr',,,

D.t AACS Sensors & Actuators

.

:lind Detector

~

mos

ct~.

~nt:erface Cont[o.l

~

~

Pyro

Bus

Fire Signals r------for Pyros

Propulsion Initiat}on Unit

iF'ire Signals Propulsion

Bart. Pwr TeLJZE ..........

....

Transponder ~

hrq

,oo,o,o,,oo l

.......................'1tl ..............~°°"'°° ................... I

....................................... l,,,,oo°,coo,,o

!'°'°"°°°

I~°o,

.........

/1 i oar~er I t I P"ckets , ! [ ~ """ [ i n ~ I/Fforins~ruments Transponder2. ~ ~ 1 MoaulaTo ~ ............. w-.{ . . . . . .

"°°°°°°°°"t-J~

............................................................. .,

....... ~ ,,ii2Li21. i,iii,iii

IL

....................................

Modulation

L

~roms/c

/

TLM

Figure 5.2 Detailed block diagram of C&DH.

144

5. Command and Data Handling Subsystem CPU BOARD

SYnc .....................

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.

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.

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.

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.

J

.....................

_i.................................................................

RAM

"I PE'~ ......................................

I/O tnterrupt~ Lower

S/C Main PWR

................................................

Voltages

CLK Swap ME M

DC/DC Converter GND

RUN

,.~

1/0 Board

cicoi! l

_m_caz

Mi

Figure 5.3 Flight processor computer block diagram.

the ~ 0 board, the memory' bonds, the DC-DC convener board, and the hm-'ness board. The CPU has its own clock and PROM. The flight processor computer performs both computational and control tasks where reliability of operation is paramount. 5.2.2

COMMAND INTERFACE CONTROL UNIT

A block diagram of the command interface control unit (CICU) is shown in Figure 5.4. The CICU is the interface between the flight processing computer and the rest of the command and data handling subsystem. The CICU receives and decodes uplink command messages and data transmission to the flight processor computer. It receives and buffers data and control signals from the flight processing computer and transmits them to the appropriate spacecraft unit. ~ e CICU also receives and buffers contmI signals from spacecraft units and transmits the priority intermp~ to the flight processor computer. Finally, the CICU provides the clock for the flight processors, the CICU itself, and other spacecraft units. As shown in Figure 5.4, the CICU assembly communicates with I/O boards, the altitude control electronic board, the timing board's uplink processor board (for command processing), the uplink formatter interface board, electronic power conditioning, and other miscellaneous logic boards.

5.2. More Detailed View of C&DH

I/0 B u s to

[pr ..............................I....P..r..o c e s s o 4 I

up~iok

I_.

J,

ETU

.

1/(3 Buffers

b~F=.tro! !

Processor TLM buffets

. . . . . . . . .

I

t/O

................................I.............. I Processor I

oce.~r I'~] ~ .

145

Buffered CPU |

Uplink Data

Up!ink Data

MasterCLK

I

1

Bus Control

| ~

!

Selected Out~ts Inpu~ RWA Torques

Buffered CPU MasterCLK ii0 !=-=...........:" : :: ":":":":":': --; ....... :::

l |

........................................................... Up,ink

,,oBo°,o

Processor

Figure 5.4

5.2.3

• " " Buffer Selection

I!o

Processor TLM buffers Data

I 1

v

~10

Buffers

Command interface control unit block diagram.

SIGNAL CONTROL UNIT

The signal control unit (SCU) accepts input signals from the CICU and conveys these signals into high-voltage, high-current interfacing outputs. These signals activate the relays on control logic contained within the SCUm switch high-level voltage and/or current interface functions which cannot be performed directly by the CICU. Figure 5.5 shows a block, diagram of the SCU. The distribution, of commands received from the CICU controls the following functions internal to the signal control unit. The distribution of commands received from the CICU controls the following functions internal m the SCU: ° Latching relay contact closures , Temporary relay contact closures of nonlatching relays • Transistorized thruster solenoid drivers ° Pyro relay drivers ° RF device relay drivers The signal control unit outputs control the following functions:

• Spacecraft battery power switching • Biprop hydrazine thruster and latch values

146

5. Command and Da~ Handling Subsystem

REA....cr~sDs HTR-cMD-s

F~.~o~

PWR

.... F

~ ....................................................... , ~' ,

I

I

.Tq o

Bi.prop, HTR CMD, l - - ~ . ~ _ i d----;~,,; .....1...3 ENABLE'

............

REA and Bi-Prop

V

l-,,J

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l R_EAValves

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REA and BiprOp"valves

.

Figure 5,5 Signal control unit block diagram. Examples of signals and driver circuits found in SCU.

• Dual t h e ~ a l control and heater power switching • ~rotechnic devices • RF device swishing drivers o Payload instoament, heaters, and data system

5.2,4

MASTER CRYSTAL OSCILLATOR

The master crystal oscillator consists of two ovenized oscillators, two buffer amplifiers, two power supplies, one switchover detector circuit and switchover logic circuit~. In normal operation all elements are powered and available for immediate use, Figure 5,6 shows a block diagram of the MXO assembly,

5.2.5

GIMBAL DRIVE ELECTRONICS

The gimbal drive electronics (GDE) units drive the gimbal ~ v e assemblies for the two solar arrays and main antennas. The GDE commands the azimuth and elevation position and performs the following functions:

5.2. More Detailed View of C&DH

DCIDC Conv.

147

Fh ,

d - F F r i m . Buffer A

Master OSC

Switchover Control

DC/DC Conv.

Primary Master OSC

Buffer B Master CLK Sec.

Figure 5.6 Master crystal oscillator block diagram.

Receive and decode a bit serial motor command word from three ClCUs Provide modes of stepper motor drive pulses Operate in both forward and reverse directions Hardware stops at the maximurn limits of operation Store shaft position data word of both azimuth and elevation actuator shafts Provide status telemet~y A block diagram of the GDE is shown in Figure 5.7 5.2.6 PYROTECHNIC RELAY ASSEMBLY

The function of the pyrotechnic relay assemblies (PRAs) is to provide command firing currents to the spacecraft electroexplosive devices (EEDs) and protect against inadvertent firing of EEDs. Each PRA consists of the following elements: primary and backup circuits. relays to deliver firing currents to EEUs, safetyrelated res~storsto bleed stray voltage. fusing resistors to preclude circuit shol-ting, and special-purpose shielded connectors. A simplified schematic diagram of a PRA circuit is shown in Figure 5.8.

148

5. Command and Data Handling Subsystem

~ [

==............ : : .................

~~~D.r!versl

'l ............................

................................

i !

t .

.

.

.

.

.

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.

.

.

.

.

.

....I'"

E!evatien Azimu~ MotcaTLM

Azimuth and E!evat~n Limi~

Data S ~ r l

Data Transfer

Figure 5.7 Gimbal drive electronics block diagram. 5.2.7

ENGINEERING TELEMETRY UNIT

A block diagram of the engineering telemet~ unit (ETU) is shown in Figure 5.9. The ETU accepts telemetry from the spacecraft in serial digital, discrete digital, and analog forms. It generates and maintains the spacecraft time code and combines spacecraft time code and telemetry for time tagging of data. It generates telemetry packets for transmission to the flight processor computer. 5,2,8

PAYLOAD DATA A S S E M B L Y

A block diagram of the payload data assembly (PDA) is shown in Figure 5.10. The PDA conveys commands received from the spacecraft to the payload assemblies, such as those found in wireless communication transponders. The PDA collects source data packets from the wireless instruments and the spacecraft. The PDA formats these data into transfer frames and provides it to the spacecraft in data streams. The PDA provides a standard interface to the wireless insmaments.

5.3

Fundamentals of Modulation Theory and Coding

Baseband signals that contain the information inside the commands to be uplinked to a spacecraft are basically analog signals that need to be digitally convened in order to properly encode info~ation in. them..

5.3. Fundamentals of Modulation Theory and Coding

Pyro Return

I1 !

149

e.turn

FIRE Command

> S/C PWR Main

Coil-Hi

EED Pyro Bus

F i b r e 5.8 Pyrotechnic relay assembly block diagram.

5.3,1

DIGITAL MODUI_ATION

Pulse code modulation (PCM) is a form of digital-to-analog conversion in which the information contained in the samples of an analog signal can be represented or shown in the form of digital words in a serial bit stream of words, ff we say that each digital word has n binary digits, then M = 2'~ unique words are possible, and each coded word has a certain amplitude level. This process is called quantizing, which means that instead of using an exact sample of the analog waveform value, the sample is substituted by the closest allowed value co~esponding to one of the code words. PCM modulation has the advantage that the circuits involved in generating such modulation are simple. Furthermore, PCM signals are very easy in timedivision multiplexing. Finally, the noise performance of a digital signal is usually superior to that of an analog signal. The main disadvantage of a PCM signal is that it uses a much wider bandwidth than an analog signal.

150 Serial CMDs

5. Command and Data Handling Subsystem Serial TLM

Seriat TLM

Sedal TLM

Digitai TLM

Anatog TLM

On/Off CMDs,

PWR

Processor Internal Voltages

Controls Signals

ADRS/Data Bus CnU

Packet I!F

PDA TLM Packets

ETU Star. Indicators

Eng, Data Timecode Data

Clocks and Sync. Pulses

Figure 5.9 Engineering telemetry unit block diagram.

The PCM output is obtained by first sampling, then quandzing, and finally encoding, as shown in Figure 5.1 I. The sampling operation first generates a pulse amplitude modulation (PAM) signal as shown in Figure 5.12. The analog waveform W(0, which is band-limited by B(z), can be converted to a PAM signal using natural sampling, giving % ( 0 = w(o s(o

(5.1)

where s(t) = ~ , n . ........

\

T

/

is a rectangular wave switching at a frequency j~ = l/T~ -> 2B(z). The spectrum of the naturally sampled PAM signal is given by its Fourier transfo~,

5.3. Fundamentals of Modulation Theory and Coding

151

Data Payload interfaces

CICU

Switch Power Supply Electronics

~

Ser, CM L___

. SCU

Discrete

[

.

.

.

.

[DC/DCConv.

: S/C CMD. CLK, t/F ..C0 ntrol ........

CMDs

teed.Solomon ncoder

TLM Module

RAM, PROM

TLM Ana. PDA on

0j-]..j. O .............. .

Contm!

J

Sw[tchPssi TLM

..................

I 1

" ! ..................................

Figure 5.10 Payload dam assembly,

W~( f ) = c .....

~

sin ~ n d w ( f _ , ~ ) , ~ nd

....

(5.2)

where c is the duty cycle given, by c = ffZs. The quantization procedure is illustrated in Figure 5.11 for the M = 8 level case. If the steps are of equal size, the quantizer is said to be u n i f o ~ . Because we are approximating the analog values using a finite number of levels, error is then introduced as pan of the output. The error waveform is shown in Ngure 5.1 I. We can minimize the channel noise by sampling at the Nyquist rate of 2B(z). Finally, the PCM signal is obtained from the quantized PAM signal; a specific code must be developed to represent a particular quantizeA level. For example, the code used in the PCM signal is called Gray code. The bandwidth of serial PCM is such that Bee M > 1/2R = 1/2 nf~,

(5.3)

where n is the number of bits in the PCM word (M = 2n), fs is the sampling rate, and: R is the bit rate.. The bandwidth of the PCM signal must be maintained

152

5. Command. and Data. Handling Subsystem

Vout

8 M=8

-8 -6 -4

'.2468 ""-4

--6 (a) ~ I

• "11"--- (Sampling Time) I

(b)

/ PAM Signal

Quantized PAM Difference between analog signal and quantized PAM signal .

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.

. PCM Word

(c)

!

.

1

1

1

O011001

10001

I

1

I

t

t 1 10101

1

(d) F~are 5.11 The PCM process.. (a) Quantizer output-input characteristic; (b)analog signal, fiat-top PAM signal, and quanti~d PAM signal.; (c) error signal, and (d) PCM signal.

5.3, Fundamentals of Modulation Theo~ and Coding

153

Imputse Train Sampling

! Ws(t)

~J

t

•

..............

...........

___t~ ~ Ts ~ 1

Figure 5.12 PAM signal representation.

significantly greater than that of the co~esponding analog signals that it represents.

5.3.2

NOISE ISSUES IN PULSE CODE MODULATION

R e analog baseband signal which is recovered from a digital-to-~alog conversion can be c o ~ p t e d by noise. There are two main sources for this noise: I. Quantizing noise that is caused by the M-step quantizer at the PCM transmitter. 2. Bit errors which are recovered in the PCM signal. These bit errors can be caused by channel noise and/or bad channel filtering, which can cause intersymbol interference (ISI). The term ISI comes from the fact that pulses corresponding to any particular bit will smear into adjacent bit slots as a result of a bad filtering such that the filtered pulse will be elongated.

154

5. Command and Data Handling Subsystem

The recovered analog peak signal-toonoise ratio is given by

S 3M 2 (N)peak = 1 + 4 i.~d2-'~'2- i) Pe"

(5.4)

The average signal power to the average noise power is (~)

M2

OUt

1 +4(M

-- I) P e

(5.5)

where M is the number of quantized levels used in the PCM system and Pe is the probability of bit error in the recovered PCM signN before it is converted back into an analog signal using the receiver DAC. Correcting bit errors reduces the value of Pc" Often, Pe can become negligible, and in such cases, the peak SkY due only to quanfizing errors (i.e., no bit error from channel noise) can be written as (5.6) peak

and the average S/N due only to quantizing errors becomes

OUt

Reducing the value of Pe is a funcdon of channel capacity. Claude Shannon showed that for the case of a signal plus Gaussian noise, a channel capacity c (bits/see) could be calculated by

where B(z) is the channel bandwidth in he~z and S/N is the signal-to-noise power ratio (in watts). Furthermore, Shannon stated that a finite value of SIN l i ~ t s only the rate of ~ansmission. ~ a t means that the probability of error Pe can approach zero, provided that the info~adon rate is less than the channel capacity. This means also that in order to approximate this scenario, coding must be used. Coding reduces Pe for the following reasons: I. The nature of redundancy: redundant bits are added by the encoder to increase the peculiar form of each transrrdtted digital message

5.3. Fundamentals of Modulation Theory and Coding

155

2, The nature of noise averaging: The coding can be designed so that. the receiver can average out the noise over long time periods Most coding can be divided into two groups: block encoding and convolutional encoding. Block codes is a mapping of K binary inputs into n output binary symbols. Block codes are memoryless. Among the most common block coding schemes are Hamming and Reed-Solomon. Convolutional encoding has memory'. The convolutional coder accepts k binary symbols and produces n binary symbols where the n output symbols are affected by v + k input signals. Memo~ is present, since v > O. Concerning quantizing noise, there are four kinds: overload noise, random noise, granular noise, arld hunting noise.

Overload noise The level of analog waveforrn at the input of the PCM encoders must be set such that its peak value does not exceed the design peak of the quantized V volts. If the peak input exceeds !< then the recovered analog waveform at the output of the PCM will have some fiat tops near the maximum values. Random noise This noise is the result of random quanfization errors in the PCM system under normal operating conditions. Granular noise

When the analog input level is reduced to a relatively small value with resp~t to the design level, the error values are not equally likely from sample to sample.

Hunting noise This occurs at the output of a PCM system. It can occur when the input analog waveform is almost constant, even when there is no signal during these conditions. 5.3.3 BINARY CODING The PCM signaling of " 1 " and " 0 " can be represented in different serial-bit signal f o ~ a t . The most popular formats are shown in Figure 5.13. There are two kinds of formats: return-to-zero (RZ) and nonretum-to-zero (NP~). For RZ coding the waveform returns to zero for about half of the bit interval, Based on the rules which are used to assign the quantized voltage to represent the binary data, the w a v e f o ~ s can be further classified:

Unipolar signal In positive unipolar logic signals, the logic 1 is represented by a positive voltage and 0 by a zero. This is often called on-a~ff keying. Polar signal The binary l's and O's are represented by positive and negative levels.

156

5. Command and Data Handling Subsystem V(t)

I !

I

v(t) t Polar N ~

0

1

I

t

i

I

I

t

i

!

l.....V...°..i.~........:.

1I

0

I I

i

I

i

1 1

11

t

I

!

!

0

I

! I

I I ............. i

t

.........................................

.Vo....(-! ..................... }i_~..t ....................... 1

t

t

I

!

I

t

I

Unipolar RZ .......................... 1 ........

I ..................

I

i

v(~

r

T~~O_~_..

Bipolar RZ

vo

V(

Manchester NRZ

J

I

I ._I I

t

h ~

i

t

.......I-I ........h......i.......................... i

I

........................

1

I

I

I

I

I

!

I

I

t._ ...................t t._............ I..i ................. li................. ~M ......II ...........-q, ............. ........... ................ I._l

I

t

I

t

i __ I

t

I

__ I I

i

I

t

hrh, r h ,

...................... i/j l--/.J

.........U

" ~

t

........ E

Figure 5.13 Binary signaling format.

Bipolar signal Binary l's are represented by alternately positive and negative values. The binm-y 0 is represented by a zero value. Manchester signal Each binary I is represented by a 0 positive halfvalue period pulse followed by a half-value negative period pulse. A b i n ~ 0 is represented by a negative half-value period pulse followed by a positive half-value ~riod pulse. The power spectral densities of these signals are as follows.

5.3. Fundamentals of Modulation Theory and Coding

157

For a polar NR~ signal, the power spectral density is given by

(

Ppolar ( f )

= r \

rr f r

(5.9)

/ '

where r is the pulse width. For a unipolar NRZ signal, the power spectral density is given by 9

i

Pun+polar(f) = 2 - r \

+fr

]

+ I

(5.10)

2 ~f)+

The main. disadvantage of unipolar N ~ is the waste of power because of the presence of a DC level, and the ~ectrum does not aoproach zero near DC. For a unipotar RZ signal, the power sp~tral density for unity power is given by 2¸

P+~(+f ) - 4

~rf v / 2

I + -T

This equation shows that the first null bandwid~ is twice that for unipo!ar or polar signaling, since the pulse width is half the width.. There is also a discrete term at f = l/r; therefore, this periodic component is used for recove~ of the clock signal. For a bipolar RZ signal, the power spectral density for a unity power signal is given, by

Pbipoiar (f)

- + [

rfr/2

'-+>+i J (2-

2 cos ~rfr).

(5.12)

Finally, for the Manchester NRZ signal, the power spectral density for a unity power signal is given by

Pma,~cheswr(f)

= r[

~fr/2

sin2

"

(5.I3)

As previously stated, intersymboI interference arises when pulses are not properly filtered as they pass through a communications system. The pulse will "spread" in time, and the pulse for each symbol will be smeared into adjacent time slots as shown in Figure 5.14.

158

5. Command and Data Handling Subsystem

lntersymbo, I Interference 1

0

1

0

........................................................................................................................

i ...........................

I

I

i

i

t

Sampling Points

F i b r e 5.14 Example of ISI on received pulses in a binary system.

5.3.4

DELTA PULSE CODE MODULATION

When consecutive sampled values of an analog input are found to be v e ~ close to the same value, there is considerable redundancy, which is not an efficient use of the bandwidth and dynamic range of PCM. One way to minimize redundant transmission and also to diminish the bandwidth of PCM signals is to transmit PCM signals corresponding to the difference in. adjacent sample value. The scheme is usually known as differential pulse code modulation (DPCM). Delta modulation (DM) is a special case of DPCM in which there are two quandzing levels. The delta modulation scheme is presented in Figure 5.15. Notice that the delta modulation system does not need an analog-to-digiml conveaer or a digital-toanalog converter. Therefore, it is much less expensive, which is its main attraction. In Figure 5..I5 a comparator is used as a subtractor and a two-level quanfizer so that the output is either + V or - V (binary). In essence the delta modulation signal is a polar signal, and a representation of the input and output w a v e f o ~ s is shown in Figure 5.t6. From the figure it can be observed that the accumulator output cannot always track the analog input signal. This quantizing noise may be classified into either slope, overload noise or granular noise. The former occurs when the step size d is too small for the accumulator output to follow fast changes in. the input signal. Granular noise

5.3. Fundamentals of Modulation Theory and Coding

Analog Input ....................................... W(t) -l .........................................i

159

PAM ~ ~ + " ' ~ "

L

]

fs

1

/

z(t)

~ ~'AC~muato ...~-................ Figure 5.15 Delta modulation system

Analog inputand Accumulator OutputWaveform GranularNoise

Slope Noise

Vc~ Y ( t ) t

-Vc i Delta Modulation Waveform Figure 5.16 Delta modulation waveforms.

160

5. Command and Data Handling Subsystem

occurs for any step size, but it is proportional to the step sized, so it is important to minimize d. An optimum value for d depends on the nature of the input signal and on the sampling frequency used. Note that if d is increased, the granular noise will. also increase, but the slope overload noise will. decrease. It can be shown that Lhe average signal-m-noise ratio out of a DM system with a sine-wave fast signal is given by S

-

3

.s

(5.I4)

ou, - 87r f ~ B '

where £ is the DM sampling frequency, fa is the frequency of the sinusoidal input, and B(z) is the bandwidth of the receiving system. Fughermore, the step size d required to prevent slope overload noise for the same sinusoidal input is given by d > 27rLA

(5.15)

where A is the magnitude of the sinusoidal input. £3.5

B A N D W I D T H COMMKNICATION FOR DIGITAL SIGNALS

A bandpass waveform has a spectral magnitude that is nonzero for a frequency region concentrated around a cartier frequency f., where fc > > 0. Elsewhere the spectral magnitude is negligible. Digital modulation, is the process by which a digital baseband signal that contains the source of information is imparted with a bandpass signal of carrier frequency ft. The bandpass signal is said to be modulated, while the baseband signal is called the modulating signal. Therefore, the power sp~tral density of a digital bandpass signal is given by !

P ( f ) = ~[em ( f - fc) + Pm ( - f -

£)],

(5.16)

where f~ is the carrier frequency and Pm (f) is the power spectral density of the mapping of the baseband signal re(t) onto the complex plane. The mapping function outlines the type of modulation. Bandpass digital modulation can be divided into two main groups, binary and multilevel types. The most common binary bandpass signaling methods (described schematically in Figure 5.17) are as follows" 1. Amplitude-shift keying (ASK) consists of switching a sinusoidaI carrier on and off with the synchronization of a unipolar binary signal

5,3, Fundamentals of Modulation Theory and Coding t

i 0

,

!

I

1

1 1

I 0

11

1

Unipolar M o d u l a t i o n ............................... lr . . . . . . 1

t 0!1

161

1

1

I

IO

I 1

t

. . . . . . . . .

t

I

I

I

I

i

1

t

t

1

I

1

I

t

t

t

t

t

I

I

1

I

t

1

Polar Modulation

I

ASK

,

....

t

?~V' 'AA ' ',5A' . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

l

'A ^'

I

i

'A A'

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

t

t

AAI~ p\IAA I

BPSK

I

I FSK ~-===~---~-=,

..:.

I

I

' I

I

i

I

I

I

I

i

t

t

I

I

1

1

1

!

I

I

Figure 5,17 Bandpass digitally modulated signals.

2. Binary phase-shift keying (BPSK) consists of shifting the phase of a sinusoidaI can-ier 0 ° or 180° in synchronization with a unipolar binary signal 3. Frequency-shift keying (FSK) consists of shifting t.he frequency of a sinusoidal carrier from a frequency representing the sending of a binary 1 to a frequency representing the sending of a binary 0; it is similar to FM carrier modulation with a binary digital signal N o . a l l y , the bandwidth of the digital signal must be limited to obtain s~ctral conservation when the information is transmitted. This can be accomplished, for

162

5. Command and Data Handling Subsystem

example, by using a premodulation filter to limit the bandwidth without causing intersymboi interference.

5.3.5.1

ASK Modulation

The ASK signal can be represented mathematically by S(t) = Am(t)cos 27rL t,

(5.17)

where m(t) is a unipolar baseband data signal as shown in Figure 5.17. The complex envelope is given by E(t) = Am(t). The power special density of this complex envelope is proportional to that of the unipol&r signal and is given by

Pro(f)=-2"

fi(f)+

\

7rfr /

'

(:5.18)

where re(t) has a peak value of V 2 so that S(t) has an average normalized power of A/2. The power spec~m density of the ASK signal is then obtained by substituting Equation (5.18) into Equation (5.16). The result for positive frequencies is shown in Figure 5.18.

I I

....f C 2 ~

-~-..................F . . . . . . . . . . . . . . . . . f o

8 (Sin[n(f-fc)k / {n(f-fc~c}) 2

................................................. ] ..................................... f c + 2 / ~

2/T Fi~

5,18 ASK bandpass digital signal spectrum.

5,3. Fundamentals of Modulation Theory and Coding

163

5.3.5.2 Binary Phase-Shift Keying The binary shift keying signal is represented by S(t ) = A cos[2n'fc + Dpm(t)],

(5.19)

where m(t)is a polar baseband data signal and Dp is the phase sensitivity of the phase modulator. The level of the pilot career is set by the value of the peak deviation 80 = Dp. For digital angle modulated signals, the modulation index k is defined by k = 2 80/zr. ~ e complex envelope for BPSK is E(t) = jAm(t).

The power spectral density for the complex envelope is given by

{Mn ~-fT"i

(5.20)

Pro(f) = A.2r,k ~fT ]'

where m(t) has a value of + 1., which means that S(O has a normalized average power of A2/2. The power spectrum density of the BPSK signal is evaluated by substituting Equation (5.20) into Equation (5.16), resulting in the spectrum of Figure 5.19.

1

~ ~ - 4

fc" 2/~

A2/4 /~!'~_

F...................................... fc ....

L~

......................... ~ f

4 (Sin[=(f-fc)]~/ {=(f-fck}. )2

-I

fc +2/~

Fibre 5.19 BPSK bandpass digital signal, spectrum.

164

5. Command and Data Handling Subsystem

5.3.5.3

Frequency-Shift K~4ng

The frequency-shift keying signal is mostly generated by feeding the data signal into a frequency modulation as shown in Figure 5.20. A FSK signal is then represented by S(t) = A cos 2 ~ f r + Df

m(~d

,

(5.21)

where m(t) is the baseband digital signal. If the data input is b i n ~ , as with a polar baseband signal, then the resulting FSK signal is called a b i n ~ FSK signal. The approximate transmission bandwidth B t for the FSK is given by Carson's rule, Bt = 2(fl + 1)B,

where fl = AF/B, B is the bandwidth of the digital modulation waveform, and i F is the peak-m-peak frequency deviation, or AF = Dfl2zr form = + 1. The power spectrum density is given by A27 2 ) 2, . P(f) = - ' ~ k ~ ( f [.1 + LI~ (/)] + k (/)[1 + Le2 (/)]

(5.22)

+ 2Lt.2 (.f)k~ (f)k 2 (f), ...................

: : . : . . . : ..: .. _ = .

,,,,

,,

Electronic

Oscillator Freq.=f1

Switch

!

............... 1

lib,-.

.......... osc,,,a,or

FSK Output

...............................

Binary Data input re(t) ......

•. . . . . . = : . . . .

=. . . . . . . . . . . . . . . .

,,,,= .................................

::::::::.:

,,==

:

: = : . = : = . . = . = . . . : . : . . . : . . : . . : : . . . . . . : ....

Discontinuous-Phase FSK Figu~ 5.20 Generating an FSK signal.

5.3. Fundamentals of Modulation Theory and Coding

165

where kn(f) :

sin(~ 1 " ( f - A F ( 2 n - 3))) T ( f - A F ( 2 n - 3))

Lnm(f) = c o s [ 2 ~ f T - 2 ~ A F T ( n + m - 3)1 - cos(27r AFT)cos[2~ A F T ( n + m - 3 ) ] 1 + cos2(2 ~ AFT) - 2~os(2 ~ AFf)cos(2 ~rf~') AF is the peak~to-peak frequency deviation, T is the binary pulse width, and the digital modulation index is given by k : 2AFt. (It is assumed that k ¢ 0,1,2.,... ; otherwise, if k = 1,2,3 . . . . . . etc., we get delta functions.) 5.3.5.4

Multisymbol Signaling

The bandwidth that is required to transmit a baseband digital sequence can be reduced by multilevel signaling: the combination of successive binary pulses to form a longer pulse requiting a smaller bandwidth for transmission. If a set of M = 2 '~ symbols is used, with n the number of successive binary digits combined to form the appropriate symbol to be transmitted, 2n bits ls!Hz can then be transmitted using the Nyquist rate. The binary rate is given, by R (bits/se~). Consider the case where n = 2, M = 22 = 4. The resultant set of four bina~ pairs ~,01,t0,11 is used to trigger a high-frequency sine wave of four possible phases, one for each binary pair. Any of the four signals can be represented by Si(t) = A cos (2 ¢rLt + 0~),

i=

. 1,2,3,4,

-T T 2 < - t < - 2"

(:5.23) ~

The possible choices of four phase angles are. Oi = O, ± 7"r!2, 7r

In any case the phases are separated by zr/2. The preceding equation can be represented by Si(t) = Ai cos 27r~,t + Bi sin 2w~t,

~'T

T

T -< t <-- 7.

The (Ai,Bi) pairs are given, corresponding to the angles (A~,B i) = (1,0), (0,1), (-1,0), ( 0 , - ! ) ~ 0, -~r/2, ~, ~/2 (a~,"i) = (1,I), (-1,1), ( - 1 , - I ) ,

( 1 , - 1 ) =- ~r/4, ~, 7 , :i.

(5.24)

166

5. Command and Data ~ n d l i n g

t

Subsystem im

lm

O Real

l

Real

O

O

Figure 5.21 QPSK signal constellation.

Transmission of this kind is called quadraphase swift keying (QPSK) with two careers in phase quadrature to one another transmitted simultaneously over the same channel. A useful way m represent the signal of Equation (5.24) is a two-dimensional diagram representing (A~,B i) as shown in Figure 5.21. A way of generating this quadrature representation is shown in. Figure 5.22.

Cos 2 r~fct

t...................................................... OSC

~ ) ~ .........~++ o.oo.+.n,?.~ .......................~........ - - - - - -~.llMSeaal4o+Paral~e!

..............................

~

Digital4o A n a l o g Converter D=F1]2 =~ s y m b o l s / s e c

. ....... ,

L t

O

0

Sm 2~fct

1 [

.................. , I+[ ....... I ,

T=2/R

I ,

T=2/R

I +

j

',J: .....~,,, .... ............. !o

+

ot

-Cos 2 rdct Sin 2~fct

Fig~Jre 5.22 Generation of a QAM signal.

t

5.3. Fundamentals of Modulation Theory and Coding 5.3.5.5

167

Quadrature Amplitude M~ulation

A more general type of multilevel signaling is generated by letting A i and B i in Equation (5.24) take on multiple values themselves. The resultant signals are called quadrature amplitude modulation (QAM) signals. These signals have multilevel amplitude modulations which are applied in an independent manner on each of the two quadrature carriers. The general QAM signal is given by

Si(t) = Ri(t)cos(2~ f,:t + 0~),

(5.25)

with the amplitude of Ri(t) and phase angle 0~ are given by the appropriate combination of (Ai, Bi). The 16-~state QAM signal set appears in Figure 5.23. For this case M = t6 = 24, n = 4, where A i and B i are each allowed to have M levels per dimension. This 16-symbol QAM can be generated using 2(4/2 = 2)bit digital-to-analog converters and quadrature balance modulators. For the case of rectangular bit signals, the power spectral density for an envelope of APSK or QAM signals is given by (sin rrf~r~ 2 Pro(f) - 2 P f r \ /,

(5.26)

where M = 2£ is the number ofpoints in. the signal map, the bit rate is R - l/r, P is the transmitted power (watts), ~ is the signal level, and .7, of course, is the

A(t) D

ii

•

•

ii

•

•

•

•

•

•

ii

•

•

•

•

Figure 5.23 The 16-State QAM signal.

B(t)

168

5. Command and Data Handling Subsystem

rectangular pulse width. The power spectral density for the complex envelope is shown in Figure 5.24. The power sp~trum density of QPSK and QAM is obtained by translating the power spectrum density of Figure 5.24 to the carrier frequency described in Equation (5.16). Notice that for C = 1, the figure gives the power spectral density of PPSK. R e null-m-null transmission bandwidth of QPSK and QAM and the swctral efficiency of QPSK and QAM signals with rectangular data pulses is given by

Bv(,f)

= 2R / t

(Hz),

(5.27)

and the spectral efficiency of QPSK and QAM signals with rectangular data pulses is given by r/= ~

(bits/sec/Hz).

(5.28)

The absolum bandwidth of any M-ary modulation signal is 1(I + r)R B =~ ,

(5.29)

where r is the roll-off factor of the filter characteristics. R e overall absolute bandwidth of the QAM signal is

BT(f)

=

(! + r ~] \

£ ] R,

(Hz)

(5.30)

and fn M r / = ( 1 + riin2

(bits/secFrtz). 0 :t13

4W~

3Rl~

2R/~

W~

(5.31)

!0 Log [(Sin nfg ~r)/ 0tf£ x)~

R/g

2Rtg

3PJg

Figure 5,24 Power spectral density of a QAM signal..

4FV~

5.3. Fundamentals of Modulation Theory and Coding

169

Consider the general baseband communication system shown in Figure 5.25. The receiver input consists of a transmitted signal S(t) that has traveled hundreds of miles through space (atmosphere + outer space). Noise is being picked up on the way and can be represented by an input Gaussian noise. For bandpass signaling, such as ASK, BPSK, and FSK, the receiver consists of a superheterodyne receiver containing a mixer, an IF amplifier, and a detector, The analog R(t) is sampled at the clock time t = to + nZ For bandpass signals c o ~ p t e d by white Gaussian noise and the matched filter reception shown in Figure 5.25, the bit error rate for the different kinds of modulation are shown in Table 5.1 for both. coherent and noncoherent detection. In the table, No is the power spectral density of the noise at the receiver input and Eb is the difference signal energy at the receiver input, that is, T

E = f[Sl(./)

-

S2(t)] 2

dr,

(5.32)

0

where T is the length of time that it takes to transmit one bit of data (binary 1) and Sl(t) and S2(t) are defined for coherent detection as follows: ASK: Sl(O = B cos(27rLt + ~, S~(t) = O,

0 < t -----T (binary 1)

0 < t <~ T (binary" 0)

BZT Eb

. . . . . 2 ........"

Noise Channel n(t)

~

2

............................

"

•

.......................................................

.............

.......~(..X ) ...................... ~

oo0.o0

ou,~o

.oa.og .............

m~,, ~--qoov,ce-

2Cos (2 rc f2 + OC )

Figure 5.25 Baseband communications system.

!!!:,i,:; /

170

5. Command and Data Handling Subsystem Comparison of Digital Signaling Methods

Table 5.1 ::

. . . . .

:

:::::

....................

========================================================== ..................

========================================================

....................

,.: .

.

.

.

.

.

.

.

.

.

:::::::::::::::::::::::::::::::

Minimum Ban@ass Signaling

Transmission Bandwidth Required ~ is the BR Rate)

ASK

R

BPSK

R

FSK

2Af + R, where 2Af = f2 " f~ is the frequency shift

DPSK

R

QPSK

~2 __~

......................................

Error P e r f o ~ n c e Coherent Detection

E ] Q[~f 2 (Noo)

Requires coherent detection

Not use~ in practice

le.._(E~¢No)

Q[ ' ~ 2 ( No E) ] ~::::::

................

-----

Noncoherent Detection

::::::::::::::::::::::::::::::::::::::::::::::::::::

.......................

=

............

Requires detection coherent =--

:

..............................

................................

,,.

BPSK: Si(0 = B cos(2~Lt + ~, $2(0 = - B cos(2~Lt + ~,

0 < t < - - T ( b i n ~ 1.) 0 < t ~ T (binary 0)

Eb = 2B2T. FSK: S l ( t ) = B cos(2~rL1 t + 0), S2(t) = B c o s ( 2 ~ £ 2 t + 0),

0
Eb = B2.T.

QPSK: S(t) = ± B cos(2~Lt + ~ - (±B)sin(27rfct + ~,

0
Here, _B factor on the cosine carder is one bit of data and the (±B) factor on the sine carrier is mother bit of dam. Finally,

5.4. Worst-Case Analysis Guidelines for Analog and Distal Design

1

O(p) - ~

e~fc(, ~p ).

171 (5.33)

The peak signal-to-noise ratio (S/N}r~ak=o~t for the analog output of an original digitally transmitted signal which has been received and decoded t~ough a PCM decoder (digital-to-analog converter) is given by peak_ou~ =

(5:.34)

~ e average 3)'N ratio is given by

= i........+......4(M22 Tip ;. ave.

Ot!t

"

(5.35)

•

5.4 Worst-Case Analysis Guidelines for Analog and Digital Design: Examples of Use in Command and Data Handling Subsystem The WCA, in conjunction with testing, provides a method for verification of perfo~ance requirements. This analysis, preferably, would be an integral p ~ of the design process of each electronic circuit. The following guidelines have been established to assist the worst-case analyst in completing the task of analyzing a given circuit. The guidelines are presented in the recommended sequence for proceeding with an analysis, which are as follows:

I. Documenta~on requirements Though documentation is an ongoing process, it is presented first in order to give the analyst a clear understanding of the final goal. Also, proceeding with the first few requirements at the s t ~ requires the analyst to become familiar with the operation of the circuit. Understanding the circuit is the first step in developing a good analysis. 2. Functional analysis The purpose of the functional analysis is to predict the perfo~ance of the circuit under the most unfavorable combination of realizable conditions. This part of the analysis is generally performed on functional "blocks" rather than individual components. The functional analysis will be reviewed in detail and must have sufficient information for the reviewer to readily verify each calculation and result.

172

5. Command and Data Handling Subsystem

Where explicit governing specifications do not exist, self-imposed or derived. requirements shall be developed and justified in the worst-case analysis. Repetitive circuits need to be analyzed only once if there are no peculiarities from one circuit to the next. Paxameters which should be considered in the perform~ce analysis of a given circuit include the variability of the part parameters as well as the variability in the circuit inputs, loads, and supply voltages.

Note: The requirement listed herein, for verification by analysis shall not relieve the designer of the. responsibility for rigorous breadboard and prototype testing. Such testing shall, be used to demonstrate that circuit performance is within the region, predicted by analysis and shall also be used as an indicator of proper analytical modeling. Testing should include., but not be limited to, performance evaluation with electronic circuit parametrics, environmental extremes, and limits of source and load reactances. Paaicular attention should also be given to noise and transients. 3. Derating criteria and stress analysis The purpose of the stress analysis is to ensure mat no components will be overstressed under the worst combination of conditions. This will increase the reliability of the circuit. R o u g h derating is an impoaant part of the functional analysis, it is combined, with: the stress analysis since derating and stress are both done on a part basis, while the functional analysis is completed on "blocks." 5.4.1 ANALYSIS TOOLS When computer programs or special-purpose software is used to generate WCA results, every' effort should be made by the analyst to aid the WCA reviewer in understanding the function of the software program and the assumptions and data used when the program was executed. In addition, models used need to be justified as the "worst case." A few of the computer programs which may be used in accomplishing this analysis are identified next.

5.4.1.1

Digi~l WCA Tools

Mentor, Cadence are pm't of an array of computer-based simulation and test generation tools, capable of the following tasks: 1~ Modeling virtually any digital electronic circuit or system 2. Stimulating the circuit model with signal patterns (waveforms) at its inputs

5.4. Worst-Case Analysis Guidelines for Analog and Digital Design

173

3. Predicting behavior of the circuit with the stimulus (either when working properly or when it has any of a wide variety of faults 4. Generating a list of test vectors (or patterns of signal inputs and expected outputs) that can be used to program an automatic tester or test the actual circuit when built Models must be shown and justified for worst-case parameter compliance. Input and output listings, can also be presented.

5.4.1.2

Analog WCA Tools

SPICE-like (most variations) programs are among several circuit simulation programs that presently are used by a substantial poaion of the electronics industry. SPICE provides a friendly interaction between the user and the program. ~ i s program will determine, the quiescent operating point of the circuit, the time domain response of the circuit, or the small-signal frequency-domain response of the circuit. SPICE contains models for the common circuit parts and is capable of simulating most electronic circuits. Unfortunately, the canned model parameters are generally typical values. ~erefore, worst-case model parameters for SPICE must Ice user generated and verified. Model parameter justification must be provided with the analysis. ~ e analysis report must contain a schematic of the simulated circuit with nodes and branches clearly identified. The input listing, output data, and appropriate data plots must also be included in the report. 5.4.2

DIGITAL CIRCUITS

All logic systems will be divided into functional blocks to allow verification of logic functions using analytical techniques (e.g., truth tables, Kamaugh maps, state tables, computer programs, timing diagrams). A functional description of each block will be included with verification of that function.

5.4.2.1

Propagation Delay

All circuits shall operate within the worst-case minimum and maximum, propagation delay limits. No race conditions may exist that can cause an undetermined or erroneous output that is propagated through the circuits. A worst-c~e timing diagram analysis (or equivalent) shall be made of all sequential circuits to determine the effects of uncertainties in switching times.

174

5. Command and Data Handling Subsystem

Factors that shall be used in determining timing margins are. applicable component specifications, including temperature, effects, power-supply voltage effects, capacitive loading effects, clock skew, and time delay, This data shall be used to construct worst-case timing diagrams so that the occurrences of related switching times may be compared. When. signals are transmitted over lines, propagation delays shall be determined and their effects used in the worst~case analysis. The same logic families and technology should be used, when ~ssible. Ho-wever, there are situations where different logic families and devices will. be intermixed in microprocessor-based systems. As a result, there can be an interface of parts having incompatible operating parameters (i.e., timing, signal levels, protocol). If such an. interface exists, the analyst should demonstrate that all parts operate properly under worst-case conditions. If speciN testing is required m guarantee performance, these tests shall be inco~orated into the piece part of unit specifications. The interface signals between all digital devices (i.e., microprocessor bit slice) shall consider capacitive loading effects and time delay.

5.4.2.2

Propagation Dday Suggestion

Since determining the exact propagation delay p~ameters is a time-consuming task, it may be more efficient to use an. overly conse~ative factor ~ for scaling that includes all delay adjustments as a "first cut" and then. refine the resultant problem areas more carefully using the techniques as outlined here.

5.4.2.3

Minimum Propagation Delay

The minimum propagation delay cannot be scaled for temperature as the maxP mum delays are. The specified minimum propagation delays shall be used for 54FXX and 5 4 ~ S X X devices. The minimum specified propagation delay for 54LSXX devices may be modified as follows: 1. Modify minimum delay m one-half of the manufacturer's data sheet typical at 25°C (convert to 50-pF load) 2. Or, modify the minimum delay to on.e-fourth of the device's full temperature specification ~Use a factor of 2 (× maximum,ra~eddelay) for standard.TTL and a factor of 4 (× maximum rated delay) for CMOS.

5.4. Worst-Case Analysis Guidelin~ for Analog and Digital Design 5.4.2.4

175

Multiple Functions within a Single Chip

Temperature testing of multi.pie functions within a single device (e.g., quad and gate) have shown a wide distribution of propagation delays. Consequently, the worst-case analysis should utilize worst-case minimum and maximum delays and not assume matched propagation delays. If matched delays are necessary, the devices must be individually screened or have laboramiaj test data to support the matching desired.

5.4.2.5

Vcc Variation

The baseline propagation delay shall be adjusted, for variations above or below the Vcc supply voltage, from those in the propagation delay specification. The device propagation delay specifications shall be examined to determine if Vcc variations are included. The co~ection factors are as follows: • LSXX and SXX devices adjust by 2%./. t VDC • ECL devices 2%/. 1 VDC • CMOS ......use manufacturer's data book ° 54F and 54ALS ........included in vendor data

5.4.2.6

Device Capacitive Loading

An analysis of the capacitive loading effects for each signal must be performed. This analysis should include the capacitance of the devices and an estimate of the printed circuit (PC) board capacitance. The following guidelines apply: LS, ALS, F Series Each pin of the device has an associated capacitance of 5 pF maximum and t .6 pF minimum. For an input which serves more than one internal (unbuffeved) function, each additional function adds 2.3 pF maximum and 0.7 pF minimum. S Series Each pin of an S device has an associated capacitance of 9.6 pF maximum and 3~2 pF minimum. For an input which serves more than one interna! function, each additional function adds 6.9 pF maximum and 2.3 pF minimum. Output capacitance of a td.-state device is as follows:

Hi-Drive Lo-Drive

ALS

F

LS

S

15 pF l 0 pF

12 pF 5 pF

12 pF 12 pF

! 2 pF 12 pF

176

5. Command and Data Handling Subsystem

5.4.2.7

Intraboard and Connector Capacitance

The capacitance of the interconnections within a printed circuit board shall be approximated by using the following equation, which is developed from a model of a line above a ground plane" CTo T = (E r / cn)[(5 / 3)(W/h) + Z7 / (log 4h / t)], where W t h Er c n

= = = = = =

trace width trace ~ickness distance from ground plane to nearest surface of trace relative dielectric constant medium 1.1811 × 10 l° inches/sec (velocity of light) 377 ohms.

Example W t h Er

= = = =

For a two-sided board with the following parameters:

.013 inches .0028 inches (1 oz plated coppeO .0569 (for a .0625-inch-thick board) 5.2,

the result is 2 pF/inch (double-layer board). This equation will also yield the following typical results (for the layer nearest to the ground plane): Six-layer-board = 5 pF/inch Seven-layer-board = 7.6 pF/inch Eight~layer-board = 12 pF/inch. The WCA shall utilize these values or show other justification (actual measurements or equivalent analysis) for other values of interboaxd capacitance. The capacitance between pins on a printed circuit board connector shall be considered to have a capacitance of 5 pE unless data can be presented in the WCA showing a different value.

5.4.2.8

Propagation Delay Correction Factor

The specified baseline maximum propagation delay is at a specific value of load capacitance, ffpically 50 pF for M38510 (MIL-STD) devices and 15 pF for most others. The calculated total load capacitance (device input capacitance, tristate

5.4. Worst-Case Analysis Guidelines tbr Analog and Digital Design

177

device output capacitance, interconnection capacitance, connector capacitance) must be less than or equal to this specified baseline load capacitance (1.5 pF or 0.50 pF). The propagation delay may be adjusted to the total calculated load capacitance by using a cogection factor, For example, the conversion of LS device propagation delay from 15 pF to 50 pF capacitive loading requires an additional 2.8 nsec to the baseline 15-pF delay. Convea specified propagation delay to mtat calculated capacitance load by using the following conversion factors: 54LS 54S 54F (standard) 54F (buffers) 54ALS (standard) 54ALS (buffers) CMOs ECL

.08 nsec/pF .06 nsec/pF .033 nsec/pF .02 nsec/pF .046 nsec/pF .023 nsec/pF Use curves in manufacturer's data book Use curves in manufacturer's data book

5.4.2.9 Propagation Delay Correction Factor Calculation When the propagation delay correction factor is not available, it can be estimated by the following technique: t = -RC

!n(1 - Vt/Vcc),

where: R Vt

- estimate of drive impedance (max) = threshold voltage of logic family (usually located in the parameter measurement area of specification) Vcc = nominal power supply voltage; solving for t / C in nsec/pF t~ C = - R

Example

ln(l - Vt / Vcc) / 1 0 ~ (nsec/pF)..

For LS devices, R can be estimated from the Io.~ (short-circuit current):

Rmax =. 5.5 W20 mA (los) = 275 ohms t / C = - 2 7 5 ln(l - 3 / 5 . 0 ) / 1 ~ = 0.0828 nsec/pK

5.4.2.10

Open Collector Calculations

A procedure for adjusting the propagation delays of open collector devices from those specified in the data sheet to actual circuit pull-up resistor and load capacitance is as follows:

178

5. Command and Data Handling Subsystem

1. Using the resistor pull-up, capacitive load, and voltages indicated in the data sheet specification., calculate the transition time of the output wave from the stable state to the transition voltage of the logic family (often located in the parameter measurement area of the specifications). The transition time equation from zero volts is t~

= -Rc

!n(l -

v~/v~)

Vt = threshold voltage of logic family "v'~c = voltage applied to pull-up resistor.. 2. Subtract the transition time (step 1) from the specified propagation delay to determine device propagation delay excluding the output circuit. 3. Using the method described in step 1, calculate the transition time using actual circuit values of pull-up resistance and load capacitance. 4. The desired propagation delay of the device under actual circuit conditions is the summation of the internal chip delay (step 2) and the actual transition time (step 3). 5. The worst-case propagation delays are determined by choosing the worstcase combination of the specifications of the device across the Vcc range and pull-up resistor values and voltages.

5.4.2.11

Line Delay

In a critical timing path, it is impo~ant to take into account the time required for signals to propagate over the interconnections. TI'L inputs connected to a signal line tend to slow the propagation, because of the input capacitance of the loads. One method of estimating the slowing effect is to treat the load capacitance as an increase in the intrinsic distributed capacitance of the line. Sample calculations show the effect to be approximately 2.9 nsec for a 12-inch line with 6 T r L loads on a PCB with 100-ohm transmission im~dance. These calculations are based on the relationship Intrinsic delay =: X/LoCo where L o and C o are the line self impedance and capacitance.

5.4.2.12

End of Life

The end-of-life propagation delay derafing is usually about 5%.

5.4.2.13

Setup and Hold

The worst-case set.up and release time for data inputs must be determined. Timing diagrams or equivalent shall be used to determine the adequacy of setup and

5.4. Worst-Case AnaLysis Guidelines for Analog and Digital Design

179

hold times with respect to clock edges. These specifications cannot be reduced for limited-temperature operation. 5.4.2.14

Decoupling

Decoupling should be provided to eliminate unwanted feedback loops and to prevent conducted noise from being transferred to. sensitive areas. Integrated circuits normally contain interstage isolation and are insensitive to low-. and midfrequency power-supply noise. For high frequencies, extemat decoupling must be provided for each integrated circuit package. Self-induced switching transients should be suppressed by adding capacitance to each printed wiring board or module. It is necessary to use capacitors that are effective at the relatively high frequency present in the switching-current spikes. For integrated circuit boards, 1-~F ceramic capacitors should be used for this purpose in order to optimize both high capacitance per size and good RF properties. Capacitor leads should be held to a minimum length and the required, capacitance should be distributed over the PC board or module. Addition of I0- to 100-#F tantalum capacitors at the power entrance to the circuit card is recommended. Additional decoupling capacitors at the power pins should be. provided, for any driving/receiving devices, buffers, multivibrators, and oscillators. 5.4.3

EXAMPLES OF ANALYSIS

Analysis of the command unit was directed reward obtaining the worst-case outputs of the circuits in order m verify, where applicable, whether or not such outputs were within ~rformance specification limits. For analog circuits, the worst-case output was obtained by considering all reasonable maximum and minimum parameter variations, power-supply tolerances, and impedance loading at the inputs and outputs. Digital circuit analysis considered worst-case delay times and transition times (if required). These times were adjusted for load capacitance, temperature effects, radiation effects, and end-of-life tolerances. For passive devices such as resistors, capacitors, and inductors, the following tolerances were used to calculate worst-case parameter variations: t. Initial pan tolerance 2. End-of-life tolerance 3. Temperature coefficients

180

5. Command and Data Handling Subsystem

For semiconductor devices, worst-case parameter variations were. based on the following: I. Initial device tolerance 2. End-of-life tolerance 3. Temperature coefficients 4. Solar radiation effects

5.4.3.1

Operating Enfironment Device Derating

The operating environment of the command interface assembly will influence the device switching ch~acteristics due to space radiation, temperature, powersupply variations, aging, and procurement measurement error. The total, environmental effect on the device switching characteristics can be obtained by an RMS of the individual components. Listed below are the individual component and .an.. estimate of their environmental ranges. The estimates are based on a + 109°C junction temperature at an ambient temperature of +75°C, radiation dose of tess than 10 k~ad (Si) (parts hard to 50 ~ a d (Si)), power supply variation of +5%, an aging factor of 12.5% (5-year mission), and a procurement measurement error of 10%" Radiation 5% Temperature 25.2% = (109°(; - 25°C) (0.3%PC) Power-supply variation 10% Aging I2.5% (5-yeax mission) Procurement 10% Total R)4S tolerance = 2

2 .......... + (i2.5%i 2' + ( i 0%) 2i + ( 5 % i 2:~= 31.9%.

This 1.32 derating factor should be applied to maximum propagation delays and maximum transition times at the actual in-circuit capacitive loads at 25°(2..

5.4.3.2

Best-Case Analysis

In performing a thorough worst-case analysis, best-case conditions must frequently be compared to worst, c ~ e conditions. For example, on a given bus the worst-case tristate, device must turn off before the best-c~e device turns on to prevent a bus fight. Similarly, worst-case data must meet flip-flop setup times for best-case clocks.

5.4. Worst-Case Analysis Guidelines for Analog and Digital Design

181

For best-case analysis, use minimum propagation delays and transition times at 25°C. If minimum values are unspecified, use one-fourth the maximum values or one-half the typical values. In many cases° best-case and worst-case signals cannot occur simultaneously and should not be compared. For example, you cannot have one gate hot while another gate in the same IC is cold. This analysis must, therefore, be applied judiciously. As a guideline, to compare worst-case skew between two signals, apply the worst-case derating factor to maximum propagation delays and transition times for one path, and apply the best-case derating factor of 1 . ~ to the minimum propagation delays and transition times for the comparison path. However, be sure to use maximum delays at 25°C for worst case, and minimum delays at 25°C for best case. 5.4.3.3

Transition Times

The time at which the output of a gate switches with respect to its input is a function of the gate's propagation delay and the previous gate's transition time. A factor, called the "K value," is, therefore, added to the delay equation to account for transition times. The K value is defined as the transition time from the output 50% voltage point to the input device one or zero voltage threshold level. Table 5.2 lists K values for various input thresholds. 5.4.3.4

Device Loading Factors

R e command interface unit board construction is a welded wire approach which adds capacitance factors to the normal input COS/MOS load capacitance. A device load is composed of the following sources of capacitance:

Table 5...2 ==========================================

........................ :::::::::::::::::::::::::::::::

::::::

=

...................

,., .,.:,

...........................

Load Input Voltage Level (Vr~D = +10 VDO

K Value

2 3 or 7 4 or 6 I or9

3/8 1/4 1/8 I/2

..............................

- ..........

;

...............................................

~

.......................

=

,,:,:,:,:,:,,,

.................................

•

.

:

.

.

.

182

5. Command and Data Handling Subsystem

1. The number of COS/MOS input loads 2. The length of wire connecting: the COSMOS loads 3. R e board, connections that a signal passes through To adjust for a higher-er (-<3.5) potting material, any load calculation error, and any wire length e~or, a + 20% factor is added to the total, load capacitance, The maximum load capacitance, Ct: for a device is N

CL = 1,2 ~

C1 + L(W) + M(P),

1

where N = number of COS/MOS input loads Ct = sum of the individual adjusted COS/MOS input capacitances which form the load L = length of wire connecting loads W = wire capacitance per inch (2.84 pF/inch) M = number of board connections that the signal passes through P = capacitance per connection per pin (5.0 pF for connection, between two boards). 5.4.3.5

Device Electrical Characteristics

The switching characteristics for a device are propagation delay (tp) and transition time (tt) vs capacitance. In calculating the operational path delay, the time to reach the minimum one (or maximum zero) voltage level must be Mded .to. the pmpagati.on delay at maximum load capacitance, Figure 5.26 is an example of an operational path. with its associated delay components for CD4000's devices, Since operations are usually synchronous to some clock, the total worst-case time delay calculated for a path must be less than the minimum clock spacing. The. switching characmristics for the devices are assumed to be selected to the maximum value when purchased. In addition to selecting the C D 4 ~ series devices to a maximum switching characteristic, a second parameter selection will be made on a CD4041 device to obtain, a tighter voltage-transfer characteristic. During the time delay calculations, if a gate drives a gate contained on the same chip, it can be assumed that the difference in the input one (or zero) voltage level will not differ by more than 1.0 V. When different chips are involved, the full input one (or zero) voltage-level range must be used.

5.4. Worst*Case Analysis Guidelines for Analog and Digital Design

183

..................................

FF1

FF2 ..

#1 OUt VTH(O)

1

I

t5

t I CLK2, V TH(O}

re,

~

t

~

', "--................................ .....f ~ t ~

l

~

I

e411~.

~1:t

Figure 5,26 Path delay model.

Since the COS~¢IOS devices are specified at two capacitance loads, a straight~line approximation will, yield intermediate ted and tt. values" Total. delay = t~. + t2. + t;~ + t 4 + t 5 + t 6 + t 7 + t 8 + t 9 + t~o + t ~ + t~2, where t values are obtained at maximum load capacitance and are defined as follows: t I, t:a, t5, t 7, t 9, tj I = transition time from the output 50% voltage point to the input device one or zero. voltage threshold level = K t for the t Driver, where K is a function of load input tODD voltage level. = propagation delay for the device at the maximum load t2, t4, t6 .... t 8 capacitance. (This is the time from input voltage threshold level to output 50% voltage point, including a t e ~ for an input rise time other than 20 nsec.)

184

5, Command and Data Hand~g Subsystem

TevEN = tp,~ (20 nsec) + t t INPUT -- 20 nsec t~o = FF2 setup time t t2 = clock skew between CYI and CK_2.

Example 1 : I / 0 Board Worst-Case Timing Analysis As an example, let us look at the I/O board of the command ~ assembly, which consists of two parallel output buffers (POBs), two serial output buffers (SOBs), and one serial input buffer (SIB), all with added cross-strap logic. Each buffer receives command signals which come from the processor interface board that is located within the command interface control unit (CICU). Each POB responds to a specified processor OPR command. Each SOB, when addressed, responds to any of four processor comm~ds: OPR reset, !PR s~tus, OPR data not last, ~ d OPR data last. Similarly, the SIB (when addressed) responds to IPR data not last, IPR status, OPR begin, and IPR data last. The signal path of each processor c o - - a n d on the board is shown in Figure 5.27. Note that all user interface circuits (ire., start, clock, data, and ~ ) are derived from heritage design and were not found to be critical. P r o c ~ u r e The nine processor commands used by the I/O board have delay paths as shown. The most critical or worst-case delay path will be further analyzed and, if it meets the worst-case criteria, will validate the others. The critical timing in the circuit of Figure 5.27 rests in whether the STROBINT signal (coming from the processor interface board) will be sufficiently long to fulfill the minimum timing requirements for the clock input of the 4035's under worst-case conditions. The relationship of these processor signals under worstcase conditions is shown in Figure 5.28. It can be seen from the timing in the figure that the user-selected A1 and A2 signals will be present for a long time compared with the STROBINT signal. The other circuits are as shown in Figure 5.29. It should be emphasized that in each circuit there are various r~uirements depending on the device input at the end of each circuit. As one example, the

POB OPR 1(2) clock input of CD4035 U87 (U73) clock inp~ of CD4035 U83 (U79) clock inp~ of CD4035 U37 (U28) clock input of CD40~ U38 (U29)

Figure 5.27 Delay path illustration for other sensitive circuits.

5.4. Worst-Case Analysis Guidelines for Analog and Distal Design t=0

185

t = 2.985 uS

........................... I V~id Ad~

'at Boar~

............................................................................................. ~ (between t=9.486 psec and 9.515 psec) STROBtNT for OPR

[~ _ _ _ [ i ] .................................. ................~........ t = 586 #sec +A t45 nsec !

.......................................................................................... .

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i

.......................................[...Data ............Valid ...... on 1/O bus .

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F,,~[

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t = 7.263 psec

Figure 5.28

SOB R eOPR s ..e . .... . _ ..t

A2 ------mA1 ---7

STROBIN1

To reset

To CD4025(U93) To CD4025(U93)

SOB tPR Status

To disable of CD4503(U85)

U94

A2 A1 SOB OPR Data Not Last

To, CD4025(U93) ~

STROBINT ~ ........ ;

"-~94L~ ~,, U94

......................... A2

.............................................................................................

u94

To CD4025(U93)--~,

l?~~"/~ il

i

To PSC input ~

F

"~

'[, ! S t

[.....S .. ........... [ [40131

~

!o0,1 .....

I4o, I

Figu~ 5.29 Path delay for the SOB OPR circuits.

186

5. Command and Data Handling Subsystem

disable input of a 4503B hex buffer must be enabled (brought low)quickly enough that the data on the I/O bus is stable when the processor latches the information during an IPR operation. If the combination of propagation, transitional, and capacitive delays pushes the occurrence of the enable such that it results in insufficient setup time, then the risk of reading bad ~ m exists. Based on the data from the worst-case analysis of the processor interface board, the earliest the processor will latch info~ation on the data bus is 4/zsec following imtial valid address time (best-case delay). "lhis poses no problem for the circuits involved in IPR operation. In the circuits of Figure 5.30, the STROBINT pulse is considered to be the most critical since its width could be as short as 441 nse~. The worst-case circuits considered here are the SOB OPR data last circuit and the SOB OPR reset circuit, since each has five gae delays and a total of 13 fanouts. Both must satisfy an 80-nsec minimum pulse-width requirement at the final output of the circuit. The SOB OPR data last circuit is further analyzed as being the worst-case circuit. The first thing to do is to determine the individual output load capacitance C L for each gate output as follows: N

C L = 1.2 ~

Cl + L(W) + M(P),

t

where N = h u m o r of COSMOS input loads Cl = sum of the individual adjusted COS/MOS input capacitances which form the load L = length of wire connecting loads (assume 4 inches/load) W = wire capacitance ~ r inch (2.84 pF/inch) M - number of board connections that the signal passes through P = capacitance per connection per pin (5.0 pF for connection between two boards). Since there are no I/O connectors involved, M = 0, and the last term of the preceding equation drops out. The following list gives each load capacitance: CL(U90) CL(U90) CL(U90) CL(U90) CL(U90)

= = = = =

1.2 1.2 1.2 1.2 1.2

[(5.5)(2) [(5.5)(3) [(5.5)(3) [(5.5)(2) [(5.5)(3)

+ + + + +

(2.84)(8) + 0] = 40..4.6 pF (2.84)(12) + 0] = 60.69 pF (2.84)(12) + 0] = 60.69 pF (2.84)(8) + 0] = 40.46 pF (2.84)(12) + 0] = 60.69 pF.

5.4. Worst-Case Analysis Guidelines for Analog and Digital Design S O B O P R Data ~ .._._.__.....

~ TO CD4025(U93~_ ~-,e,~To C D 4 0 2 5 ( U 9 3 ) (

187

To C D 4 0 0 2 ( U 7 5 ) ... __.

of 0 D 4 5 0 3 ~27(U45)

..................... of C D 4 0 2 7

To K A Input of C D 4 0 1 9 ( U 3 5 )

To Disable ~nput of C 0 4 5 0 3 ( U 6 9 )

~ | B {PR B E G I N

To reset o t C D 4 0 2 7 B ~027(U45) •t of C D 4 0 2 7 B

A2 A1

(304027

3D402'7 To Pt-4B

Figure 5.30

Path delay for the SIB circuit.

Based on these C L values, the corresponding transitional and propagational delays may be obtained as follows (in nanoseconds)" tpI~IL tpLlt tWilL tTLn U90(CD4001 B)

U84(CD4011B) U93(CD4025B) U66(CD4~l B) U84(CD4011B)

57 65 65 57 65

57 65 65 57 65

45 55 55 45 55

45 55 55 45 55

188

5. Command and Data Handling Subsystem

This data may now be directly applied m the delay waveforms as outlined. Figure 5.31 shows the worst-case initial edge, and Figure. 5.32 shows best-case trailing edge. For each waveform in these two figures, the propagation, delay (tp) and the transition delay were evaluated as follows (evaluated at maximum load capacitance Q for WCA edge). Propagation delay is the time from input voltage threshold level to. the output 50% voltage point, including a term for an input rise time: tp = tp. L (or tpLH) + ((tT. L (or tTLH) -- 20 nsec)/4)

145 nsec tolerance from flight compter b ~ r d STROBINT (input).

...........~ - t p H L I .

= 57 +(45-20)/4 =64

L H ug0 output (CD4001B) ' ~"- ~ ""=-~ .................................................... ~

114(45)=12

' t PHL = 65+(55"20)/4=74 l"q["~

U93 output (CD4025B) . ~ " i .............................................................................I " ~ t

~LU = t/4(55.)=14

U660utput (CD4~iB} ........................." - \ ,

,

1 U84 output (CD4011 B) ~ Total worst case edge deIay= 561 nsec

Figure 5.31 Worst.case initial edge analysis.

tTL H = 1/4(55)= 14

5.4. Worst.Case Analysis Guidelines for Analog and Digital Design 586 nsec (actual pulse width) /

~ t

I

•

~

tpHL = 30+(25-20)/4=31

--I~~~

tTHL=tt4(25)=6

- ~ i ~" ~

; ~ c = 30+(25-20//,4•=31

i

U84 output (CD40i 1B)

,4/'

........................................................................../t ~ ~

tTLH = t/4(55)=6

J I

.

...............

~ ~ tpHL=30+(25"20)/4=31 U93 O~dtP]Jl~(GD4025B)...................."\\\(. i

i

~

~HE~~) =6

!

~ 14/I .....V.. / i

U66 outp.u

t

! i

tPLH=30+(25"20)/4=31

~ _ _ tTLH- t/4(25)=6 ~. I

U84 output (CD40118) Tota~worst case edge delay = ~ I nsec ~ 1

~tTHL

Figure 5.32 Best-case trailing edge delay.

=t14(25)=6

189

190

5. Command and Data Handling Subsystem

Transition delay is the time from the 50% output voltage point to the input device one or zero voltage threshold level. tt = K tvl.m (or tTL.), where K is a function of the load input voltage leveI as follows:

Load Input Voltage Level (Vcc = I0 V) 2 3 or 7 4 or6 I or 9

K Value 3/8 I/4 t/8 1/2

The value for K is equal to !/4 to specify a maximum logic 0 to be 3.0 V and a minimum logic 1 to be 7.0 V, The best-case edge was considered to be one-half of the typical propagation and transition delays. R e sum of the best-case trailing edge delays is 771 nsec, and the sum of the worst-case initial eAge delays is 561 nsec, leaving a worst-case pulse width of 210 nsec. This i.s further derated by 32% because of operating environment derating, which leaves an absolute worst case of 141-nsec pulse width. This is clearly greater than the. 80-nsec minimum pulse width, and shows no worst-case timing problem by a 61-nsec margin. Conclusion

Example 2: Uplink Processor Worst-Case Analysis The uplink processor function is accomplished in the CICU using two circuit boards per side. These are the uplink processor boards and the uplink extension,~TU interface board. For the purpose of this worst-case analysis, these boards will be treated as one unit. The uplink extension board also contains a separate function, the serial output buffer used by the engineering data formatter (EDF). This is a standard circuit w~ch is identical to the SOBs residing on the Type I/O boards in the CICU; therefore, a worst-case analysis of this circuit is given in the Type L/O board analysis. Each of the two uplink processors is dedicated to one of the flight computers, so this function cannot be cross-strapped. A failure of one uplink processor will disable the ground link of its respective flight computer. Each up!ink processor receives ground commands and data via the command data unit. When both CDUs are operating (two for redundancy), uplink processor 1 gets data from CDU 1, and uplink processor 2 gets data from CDU 2~ If one of

5.4. Worst-Case Analysis Guidelines for Analog and Digital Design

191

the CDUs/~ails, logic is provided to allow both uptink processors to automatically receive data from the. CDU that is still operating. In addition, each uplink processor has an independent serial input to be used by ground support equipment in lieu of the CDU input. The uplink processor receives serial data from the CDU and p e t r o l s an error correction and detection algorithm on this data. If the data passes the e~or correction/detection, it is then examined by the uplink processor to determine which type of command has been sent. This data is then made available to the spacecraft main processor, or in the case of a CICU hardware, decoded command, is decoded by the CICU to set or reset various control signals in the C&DH system. The uplink processor uses several long shift registers to accomplish its function. The rise time of the clocks used to operate these shift registers must be analyzed to determine if proper shift register operation can be guaranteed using worstcase parameters. Uplink command data rates are relatively slow (5 kHz)and therefore do not cause any difficult timing problems. However, after a 56-bit uplinked data frame has been received, e~vor correction must t ~ e place on. each of the 56 bits before the first bit of the next frame can be processed. To do this, the sequencers which control the uplink processor are clocked at 360 kHz. Cegain signals that are used by the sequencers must propagate through several logic gates during one half-cycle of the sequencer clock. A worst-case analysis has revealed that two of the signals used by the exit sequencer would not propagate fast enough in one-half of a clock cycle. The attached worst-case analysis is based on the circuit in Figure 5.33 and verifies that: this circuit design will be reliable assuming worst-case propagation delays and transition times. The interface to the B IO~ 1(2) bus by the uplink processor uses a CD4503 tristate device.

U4t (0D4027)

U43(CD4028) U24(CD4049)

U2(CD4025) U25(CD4011)

U13(CD4027)

U101(CD4012)

360KHzl

[

SEQCLK

U32(CD4041)

Figure 5.33 Exit sequencer worst-case analysis delay path.

192

5. Command and Data Handling Subsystem

This circuit is used by other functions connected to this bus in the CICU, and a worst-case discussion is given in the flight computer interface board worst-case analysis. We have to show that K input to final CD4027 flip-flop is set up in less than one cycle of the 360-kHz clock, that is, (time delays) < 1/360 kHz = 2778 nsec, In this particular example, all devices switching characteristics are derated 32% (extra margin needed) and all. capacitive loads are multiplied by a factor of 1.2. (See Table 5.3.) Total path delay can be calculated according to the reference of using K = for each IC: Total path delay = ~(5 ) {for U32 } + (130 + ~J-1

I~

+ (1.60 + T + (.109 + T -

6.

5) {for U41 }

.

80

5) {for U43} + (65 + - 2 - 5) {for U24} 6O

5) {for U2} + ( t ~ + (I~

+ T-

+ ~-

5) {for U25}

5) {for UI01} = 844 nsec.

Add to this worst-case setup time of UI3 (75 nsec) and derate 32%. The maximum worst-case delay is thus 1.32(844 + 75) = 1231 nse~. We can see that 1213 nsec <2770 nsec, so there is no timing problem in the exit sequencer.

Table 5.3

U32 U41 U43 u24 u2 I]25. U1O1 (CD4041) ~D4027) (CD4028) (CD4049) (CD4025) (CD401!) (CD4012)

Device

_._

___

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Load Capacitance (pF) Max, Propagation. Delay @ 25°C (nsec) Max, Transition Delay @ 25°C (nsec) ~

:

.

.

,

Exit Sequencer Worst-case Delay Path Calculations

.....

:,::~_.=.: ..................

;;JJJJ,..a

20~

<50

<50

<50

15

t5

t5

--

130

t6.0

65

1~

100

100

ICYO

100

60

60

56 .............

: ..............

. : .......................

:.........

.......... - - - - - :

,.

80 :: ....................................

:::::::::::::::::::::

60 .......... ::::::::......:::::::::::::.

===========================================================================================================

aU32 uses two piggy-backed CD4041 drivers. Transition time is based on one-half the actual toad capacitance.

5.5. Noise Issues in Satellite TelecommunicationsSubsystems

193

U 14 is activated on the falling edge of SEQCLK. U 103 must have its J input settled before the rising edge of SEQCLK. Signal must propagate in one half~ cycle of SEQCLK~ (See Figure 5.34 and Table 5.4.) (time delays) < 0.5/360 kHz = 1.390 nsec 1 5 {U32} + (160 + -L~) Total path delay = ~(6) - -5) + (165 + ~ + (100 + ~ -

5){U24} + (100 + ~ -

5) {U7} + (i00 + ~ -

{UI4} 5){U6}

5) {U15} - 669 nsec.

Add to this worst-case setup time of U103 (75 nsec), and derate 32%. The maximum worst-case delay is 1.32(669 + 75) = 982 nsec. Therefore, as 982 nsec < < 1390 nsec, there are no timing problems in the interrupt sequencer.

5.5

Noise Issues in Satellite Telecommunications Subsystems

The h e ~ of the satellite's mission in wireless communications lies within the telecommunication subsystem. We first examine some of the fundamentals of telecommunications link analysis, signal-to-noise ratios and thermal noise concerns. The concepts of TDMA wilt also be addressed, "We then address more complex issues about noise concerns in the atmosphere and outer space which directly affect the performance of not only the telecommunications subsystems of satellites, but spacecraft reliability itself.

U14.(CD4028) U24(CD4.049)

U6(CD4025) UT(CD4073)

U103(CD4027)

U15(CD4075)

360KHzi "-..,.

t

SEQCLK

U32(CD4~1 )

Figure 5.34 Interrupt sequencer worst-case delay path.

194

5. Command and Data Handling Subsystem

Table 5,4 Interrupt Sequencer Worst-c~e Delay Path Calculations Device

U32 U14 U24 U6 U7 U15 (CD4041) (CD4028) (CD4049) (CD4025) (CD4073) (CD4075)

Load Capacitance (pF)

200~

<50

20

15

15

15

Max. Propagation Delay @ 25°C (nsec)

--

160

65

100

1~

100

Max. Transition Delay @ 25°C (nsec)

56

1~

80

60

60

...............................................................................................................................................................................................................

:. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

========================================= ......... ,,,J,J,...

=;===

60 .::

: ..,-

.r,,,=.,,:.,,.

"U32 uses two piggy-backedCD404I drivers, Transition time is based on one-half the actual load capacitance,

5,5.1

THE U ~ q K / D O W N L I N K MODELS FOR SATELLITE COMMUNICATIONS

Consider the block diagram of Figure 5.35, which represents an uplink model (i.e., dam transmission from ground stations to satellites). The input date on baseband signal m(t), which is a PCM signal, is b~dpass using a modulation scheme (e.g., QPSK). The resulting signal is then u~onverted, bandpass limited, and transmitteA using a solid-state power amplifier (SSPA) for boosting the signal strength before it is trans~tted to the satellite. ~,rlaen the RF signal is transmitted ~ o u g h space using a high-gain ~mnna, atmospheric and space losses will experienced by the signal up to ~ e point when it is received by the satellite transponder. Within the satellite, the signal is filtered because of the possible noise absorbed in the atmosphere or outer space. A preamplifier is used to boost the received signal strength (.because of space loss). The boosted and noise-clean received signal is frequency translated for getting ready to downlink. The noise is filtered again (because of noise induced by local oscillators) and the RF power is boosted again before the signal is transmitted to each. Within the satellite receiver system, the output s i s a l can be multiplexeA to a given transponder (a transponder will contain a filter and equalizer, a amplifier, and an output filter) before it is transmitted to earth. (See Figure 5.36.)

5.5. Noise Issues in Satellite Telecommunications Subsystems

195

Upconverter

t

'.

........................ '

L(4~ R t L ,2 '4r~ ~ R t c '2 ts = . up r = t . up ,up F

/ /

......

" / f N

L -' • other

e (N up)

~oss

.........................

Other Losses ....~ ................... Pup

t satemte ............ i Transponder

L,~=c_e}yer ....... Figure. 5.35

Gup'rec

Uplink model of a satellite transponder.

f

Frequency Translator

.~.~, t

I

t

l]_o-I

"~,,

etr

i

r- -~-1

'

',.- ~

t

I

................................. t! . _ ~ _

..,~;~._.I

Rec-e.iver I "°'°°°''

Power at the Transponder Output

~'~~-'---

! J

Transmitter

Receiver

.............................

1

L=.._

Figure 5,36

, l~.x[ '1 ........

[

;=I t

= = = = _ =

.._=.=__.

:. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

.. . . . . . . . . . . . . . . . . . . .

Satellite transponder model.

, ...................................................

j_

I

196

5. Command and Data Handling Subsystem

In the return path, the transponder signal will again receive atmosphere losses and space losses, and noise will also be added to the received signal on earth. The received signal is filtered again (to remove received noise), amplified again (to boost R_r~signal as a result of space losses), and downconverted in frequency (it was frequency translated within the satellite to an optimum frequency for downlink) before the bandpass signal is demodulated and the final data output is obtained. 5.5.2

RECEIVEDPOWER

l~t us consider Figure 5.35, From the figure, it can be observed that the overall received power at the sa~llite's input is given by PR (satellite) = PTGT Gs..~-a.t:u.Plin--~k.

(5.36)

It is well kmown that PT = PEtRP

GT '

(5.37)

where PE~RP is the effective radiated power. The term Gx is defined as antenna gain for a given antenna X (X = T for transmitting antenna): Maximum radiating intensity Gx = Rad~afionintensity..of::iso~0pic...anten~na .

(5.38)

The power density (W/m 2) of an isotropic ante.nna at a distance d from the antenna is given by p(-gc,/m2) _ PEIRP 4,rrd 2.

(5.39)

Often, instead of power density, the strength of the electromagnetic field in volts/meter is given. ~ e power density is given by P(W/m) =

E (V/m) 37--'--~'

(5.40)

where 377 is the free space intrinsic impedance. A useful formula for Gx is Gx = 47rAe A2 , where Ae is the effective area of the antenna.

5.5. Noise Issues in Satellite Telecommunications Subsystems

197

An approximate expression for the gain of a parabolic antenna, which is the most widely used antenna in samllite communications (especially for ground antennas) is 4 rrf2(nz) Ae

G-

r I ...................c 2

(5.41)

,

where 7/ d ,t c f A.e

= = = = = =

antenna efficiency (---0.55) antenna diameter wavelength (m) Z99 x l0 s m/sec d3, (carrier frequency) d2,rr 4 -~ aperture area of transmitting antenna.

Actually, if the frequency is in gigaheaz and d in meters, and r / = 0.55, the preceding fo~u.la reduces to G = 60.7f2d 2.

(5.42)

Equation (5.36) can be represented as

PR (satellite) = PT GT

(4+.)

Gse~-uplink,

(5.43)

the free @ace loss (Lfs) is defined as "9

L~,:s =

•

\

c

/

fup is the uplink frequency, and R is the distance traveled in space. This distance is also given by R - [(h + re) 2 + r~ - 2r c (h + re)cos 0] ~/2,

(5.44)

where h, re, and 0 are defined in Figure 5.37. In terms of decibels, ~ u a t i o n (5.43) can be represented by the uplink equation PR(satellite in dB) = PT(dBW) - Lf~(dB) + GT(dB) + Gsatoup~ink(dB). (5.45)

198

5. Command and Data Handling Subsystem

Satellite

h

R Earth Station Site .,,

i

C B re

re

O

Earth Center

Figure 5,37 Slant range, central, nadir, and elevation angles of a geostationary satellite in space.

Using the same procedure, the power received by the e~th station in the downlink path is Pr~(earth - station) = PT~at(dBW) - Li~(dB) + GT_~a~(dB) + Gground(dB), where

Lfs(dB) =. 2O log (4Crjd°wnR)2. faown is the downlink frequency.

(5.46)

5.5. Noise Issues in Satellite Telecommunications Subsystems 5.5.3

199

NOISE T E M P E R A T U R E A N D NO~SE FIGURE

The power of noise generated by microwave and RF equipment, as well as the power generated within transmission systems, is often expressed in t e ~ s of noise figure (NF). Noise figures of communication, assemblies are impo~ant in link budget evaluations. In low-noise sources, the equivalent noise temperature (Te) provides a more practical system parameter. For a resistive element, the thermal noise power spectrum can be given by P ( f ) = 2 RKT,

(5.47)

where K = 1.38 X 10 -23 J/K is Boltzmann's constant, R is the value of the resistor in ohms, and T is the resist.or temperature in kelvins (K)(273 + °C). Another useful, te.~ is the available noise power, which is the maximum power that can be drawn from a thermal noise source: Pa = KTB(z).

(5.48)

lit does not depend, on R. The available noise power from a source can be specified by a number called the noise temperature:

e~

T = KB(z)"

(5.49)

Again T is given in kelvins. For a linear device of gain G (F) and available output power spectral density Pout (F), as shown in Figure 5.38, and assuming that the gain is constant over

V

n(t) Figure 5.38 Noise model of linear device.

200

5. Command and Data Handling Subsystem

the frequency range (j~ - B/2) <-fc <- (fc + B/2), wherefc is the center frequency within the given width, the noise figure is given by F =

Pou : KT~fl(Hz)G(F )

(5.50)

and TO = 290 K. The average effective noise temperature Te is

1),

To=To(For

N r ~ . = 10 log

I +

To (~] Ti,iK) J

(5.5~)

Finally, the noise density is the noise power that is present in a normalized l-Hz bandwidth,

N = k T e, where k is Boltzmann's constant (-1.98.6 dBmJK). If in a communication system we have a series of linear, devices (e.g., P~ preamp, transmission lines, up/down converter, IF amplifiers) c~caded together as shown in Ngure 5.39, then the overall noise f i ~ r e is given by

F _ F~ + F ~ - 1 + G1

+...+

F,,- 1

GG2

and the overall effective input-noise temperature for cascaded linear devices is given by Te- L +

5.5.4

+

G1G2

+'"

+

Tn . GIG2 .... Gn-1

(5.52)

DOWNLINK AND UPLINK EQUATIONS

The basic uplink (uplink) equation of carrier (C) to noise (N) ratio is

(5.53)

\/Vuplk/ Kup!k KTsat'

__ 2emb'Y 2 .....................................................

IF1

62

~F2

/Assembly- - - -----

O

~

n -

• ~Fn_____

Figure 5.39 Cascaded telecommunications devices for calculating noise figure.

5.5. Noise Issues in Satellite Telecommunications Subsystems

201

where Puplk

PT GT asat~.up!k ,~2plk (4~ Rupik) 2

This equation can be expressed as = 10 log

PvGT -

20 log \[ ~ ~ J \ aupIk /

+ 101og~Tsar

(5,54)

101ogK,

where Tsat is the satellite noise temperature. The downtink (dwlk) model can be represented by (~oCt)

dB~downlink

= t0 log (PT-.satGT-sat) -- 20 log 4~Rdw~k " ~dw|k + I01og~rdwlk

(5.55)

101ogK.

where aground, is the gain of the ground antenna. The overall total c ~ e r powerto-noise ratio (C/NouO at the receive earth station is given by I

Chapter 6

Noise Representations in Transponders and Multiple Access

6.0 Imroducfion At ~ e h e ~ of a satellite communications system is the transponder. The transponder consists of input and output filters, up and down converters, phase-locked loops, and traveling wave tube amplifiers ( T ~ A s . ) More modem transponders systems are using solid state power amplifiers (SSPAs)~ We now consider the nonlinear behavior of the transponder. A block diagram representation of a typical transponder was shown in Figure 5.36. Let the input of the transponder be represented by Si(t) = A cos(wet + f l ~ ) ,

(6.1)

where oJc = 2~rfc is the angular career frequency and phase of the input signal. The transponder output can ~ represented as

Sou.t = g(A)cos(wct + A ~ + f(A)).

(6.2)

If g(A) and f(A) are independent of oJc, let Ak(O, Wc + ~oj,, and flOk(t) ) denote the envelope, the angular carrier frequency, and the phase of the kth carrier, wc is the midband frequency or center frequency, which can take any value within a transponder bandwidth. For m number of modulated c ~ e r s , access to a Nansponder input can be represented by

Si(t) = ~ Ak(t) cos{(coc + oJ~)t +fk(O(t)}-

-

Ak(t) sin(o4t +fk(~O) .

Ak(t)COS(Wkt +fk(O(t)) COS coot

sin wd (6.3)

.

= X(O cos o)ot - Y(t) sin wet = ~ . ~ + y2 cos Wct+ tan- 1

202

6.0. Introduction

203

The co~esponding transponder output is

{

= Re g(~,~2 + y 2 ) . e x p

(,v/X2 + y2)

•

]

(X + jg) exp(jo4t )

}

(6.4)

Define the double Fourier t r a n s f o ~ . .

) . \ ~ (X + jY) exp[~

exp ... (

r 2)

juX - jvY] dx dy,

which means

27r Therefore,

After further mathematical manipulations,

{

&re(t) = R e exp(jo4t )

kkk

...

~Kl(~,t + f(Sl) + jKe(w2t + f(02)] ) +JKn(wnt + f( On) • N(k) ,

exPL + " " where K~, K2 . . . . and

N(k)

, K~ can be zero or any integers either positive or negative,

exp[jf ( ~ ) ] ( X

(2 7r)2 ) _

e=1

Ke tan-i

× exp £=1

+ jD H J K ~ ( A e ~ )

exp[

- j ~ - jvy]&dy du dv,

(6.5)

204

6. Noise Representations in Transponders and• Multiple Access

where .Ix is the Kth-order Bessel function. Using the polar coordinate• transformation X = p c o s (, Y=psin(,

u = ysin: r/ v= ycosr/

and performing the integration, on .£ and r/ simplifies the: expression to the following two cases. For K l + K 2 + . . . + K , , - 1,

= ff 0o

p g

" Jl(Y P) dT dp

C=1

and for K l + K 2 + . . . + K,, ~ 1, N(k) = O. Finally, the output of the transponder can be expressed as Sou t (it) = Re{N(k)exp[j(~ t + f{"(#)]},

(6.6)

where m

f~(O) = ~

Ktfe(O) m

~1

In the case of numerical computation, ~e. factor g(p)exp[ff(p)] can be approximated by L

g(p)exp[jf (p)] = ~ beJ~(oeCp), where L is the number of coefficients needed, J! is the first-order Bessel function, and a = 2~(period of the Fourier series). For a given transponder, the characteristics g(p) and tip) are known. Since g(p) and f(~) are given, the coefficients be can be obtained by an approximation, and N(k) reduces to L

tn

~=1

#=1

(6,7) which outlines the amplitude for each input signal. The be are determined by best fit from the input data of g(p) andfip) in terms of least-square error. Computer programs can calculate Sout(t) when Si(t ), g(~, and tip) are given.

6.1. Traveling Wave Tube Amplifiers in Satellite Transponders

6.1

205

Traveling Wave ~ b e Amplifiers in Satellite Transponders

The traveling wave tube amplifier ( ~ T A ) together with the klystron are known as linear-beam types of microwave device.s. Two type of TWTAs, the helix (for broadband applications) and the coupled cavity (for high-power applications) are predominant. Such devices are used for frequencies ranging from below 1 GHz and with power ranging from watts to megawatts, In satellite systems the TWTA is used as the final amplifier before the signal is transmitted. It was Lindbland [4] who first described the helix traveling wave amplifiers and was the first to explain that a synchronous interaction between an electron beam and the RF wave on a helix could produce amplification of a signal on. the helix. As previously stated, there are two basic types of traveling wave tubes. The. helix TWTA shown, in Figure 6.1 is used mostly in satellite communications with RF power ranging from tens to hundreds of watts and also for broadband applications. The coupled-cavity TWTA is used mostly in radar applications, since it is capable of providing several megawatts of power, but at the expense of a very limited bandwidth. Let us assume that a transmission line is bent into a helix, as shown in Figure 6.2. An RF signal that is applied to the left end of the helix will travel at the speed of light in a helical path along the length of the conductor. The velocity in the axial direction, the x direction in Figure 6.2, will be the velocity through the helix re.ducexl by the helix pitch. The polarity of the signal wilt alternate every half, wavelength along the helix conductor. As shown in Figure 6.2, the electric field lines extend from regions of positive charge to regions of negative

Magnetic F~using Fie{d RF Output

.

iL j

L

~

............................................. ~

~ Etectron Gun

\

~ E~ectron Beam

Figure 6.1

Hetix SJow-Wa.ve Circuit

Description of a helix ~ T

Collector

amplifier,

y

206

6. Noise Representations in Transponders and Multiple Access

Z

Figure 6.2 Electron velocity and direction through, the TWTA helix.

charge. Fuahermore, there, is also an electric field inside the helix with large axial components. When an. electron beam (generated by the electron gun. in Figure 6.1) is injected along the axis of the helix, the axial electric field components accelerate some electrons and will also slow down other electrons, in Figure 6.2, the forces on the electrons would be toward the regions denoted by 1 and away from the regions denoted by 2. The field pattern will vary sinusoidally in the axial direction. If the axial velocity of the electric field and the electron beam is the same, the electrons will experience a continuous force toward region 1 as the electron, beam goes through the helix. The electrons will start bunching in region 1. The field produced by the bunching electrons in the beam will make the electrons, on the helix to move away from the region near 1 and toward the 2 regions. This causes the field on the helix to change in two ways: 1.. The electron current flowing to the !eft on the helix from region 1 is that of cu~ent flowing to the right. This current, causes a positive voltage region on the helix m the left of region 1. In the same way the electron current flowing to the fight on the helix from mgi.on 1 produces a negative voltage to the right side of 1. 2. When the beam-wave interaction continues, the induced voltage waveform becomes much larger than the input wavefo~. When the voltage wa.vefo~ on the helix shifts to the left, decelerating field regions start moving into the electron bunching regions; energy is extracted from. the decelerating bunches, and then transferred to the circuit field, which produces

6.1. Traveling Wave Tube Amplifiers in Satellite Transponders

207

amplification of that field. Figure 6.3 shows the relationship between the electron beam and the growth of circuit voltage. There are basically three major sources of noise in ~ T A : shot noise, velocity noise, and thermal noise. Shot noise results from the discrete nature of the electron as a result of the fact that etec.tron emission from the cathode is a random process.. "velocity noise results from the wide Maxwellian distribution in velocities of electrons emitted from the cathode. Thermal agitation noise is present because of the RF circuits, which: have loss. In addition to these ever-present noise sources, there are other sources of noise that depend mainly on the design and construction of a TWTA. Most of these noise sources can be diminished greatly by good design practices and manufacturing procedures. Some of these noise sources come as a result of partition effects, flicker effects, collision ionization, secondary and reflected electrons, lens effects, noise growth, poor insulation, yawing insulator charges, microphonics resulting: from vibration, multipactors, and power-supply-induced noise. The most important noise in a TWTA is produced by the cathode shot and velocity. This noise is modified as the electron beam travels through the electron gun and into the RF region. An important factor to consider is flaat the noise on the electron beam propagates like any other RF signal on the electron k a m . Noise travels in fast and slow space-charge waves. The two sources of this noise

Circuit voltage

,\,

x,,,

,

,

,/

I /

1 i

/4"{

-"',

decelerating field

Electrondensity

/ , , ,

..-x.., , I

I .4,

1 J~k

t.,), t f i 'k ' / i \

\, ),

~,

t Jt~ I 1 f /1

"V : V ?2

!: i

distance along circuit in half wavelength

Figure 6.3 Voltageand charge density buildup in a TWTA.

208

6. Noise Representations in ~ansponders and Multiple Access

Electron Gun

..................................................................................

I Cathode I Surface I

[

t

= ..................................

f Space Chargel Low Velocity i Minimum I Correlation

I !

Drift , [" IN°iseli,

, I [ t\

"

~-#.du£1=!'!#~ t

-.

..................

~

l High Voltage I Acceleration

i I

V

'~~ i Shot Current; t

.....................................................................................................................

I Amptification 1

t t

. i~ i,.~ ~ Impedance ~Transform I,

J l l

I

'~'~

"

.............................................

RF SECTION

i I

~-- . . . . l,

11 I

Figure 6.4 Noise regions in a TWTA electron gun. we shot noise and velocity noise, and since they are uncorrelated, each source produces fast and slow waves. As these noise waves travel along the beam, noise current and velocity standing waves axe independently produced. When there is no correlation between current and velocity noise, the minimum noise figure for a TWTA is around 6.5 dB, with the correlation noise figure "below 3 dB. It has also been shown that a large axial magnetic field in the acceleration region at the input section of the RF structure is capable of inhibiting noise. In assessing the noise sources in a TWTA, Figure 6.4 shows the four noise regions into which the electron gun can be divided. The figure also shows the RF section noise regions. In the cathode region and cathode surface, shot and velocity noise are the main noise sources. In the space-charge minimum region, low-velocity electrons are returned to the cathode by the potential of the spacecharge region. The amount by which the potential is depressed fluctuates as the cathode current fluctuates because of shot noise. In a low-noise TWTA, electrons drift at low velocities for a large distance in the low-velocity region. ~ e n the electron beam leaves the low-velocity correlation region, it undergoes acceleration in the high-voltage acceleration region. Space-charge waves are sent and a standing wave pattern is produced. ~ i s sudden change in impedance produces an increase in standing wave ratio, and this causes an increase in the noise figure for a low-noise TWTA.

6.2

Distortions in TWTAs

The power output vs power input of a ~÷¢TA is mostly linear, and signal distortions are small. However, as the TWTA is driven into saturation, the transform

6.2. Distortions in TWTAs

209

function becomes nonlinear, and as a result distortions do occur in amptitude~ modulated (AM) or phase-modulated (PM) signals. These distortions occur for AM/AM conversion, AM/PM conversion, harmonic generation, and intermodula~ tion distortions. The relationship between RF power output and RF power input for a TWTA is shown in Figure 6.5. Gain is defined as the rate of output RF power to input RF power. There are basically two types of signal gains measured: the saturated gain and small-signal gain. AM/AM conversion is a measure of output RF power that results fi'om a change in RF input power. It can also be calculated as the slope of the curve of 1~ power output, vs RF input power of Figure 6.5. It is therefore often the case that power levels must be reduced because of saturation. AM/PM conversion is a measure of the change in TWTA phase length resulting from a change in RF drive level. As the drive RF power is increased, more power is extracted from the electron beam and the velocity of the beam is reduced. As the beam vel~ity decreases, the velocity of the RF wave is reduced, and this increases the phase length of the TWTA. ~ . e typical Figure 6.6 shows the variation of phase length with RF input power. As the drive power is increased, approaching saturation, the rate of phase change increases rapidly and then decreases as the TWTA saturates. As before, the slope of the phase length curve in degrees/dB is the AM/PM conversion.

RF Output Power

t

Saturated Region

Linear Region

Slope = AM/PM Conversion

RF Input Power Figure 6.5 Linear and saturated regions in a TWTA,

210

6. Noise Representations in Transponders and Multiple Access

k Phase Length Slope = AM/PM conversion

RF Power Input Figure 6.6 Power output and phase length variation in TWTA for a given input power.

As before, a maximum allowed value of AM/PM conversion is usually specified (3-10 dB below the input power that causes saturation).

6.2.1

1NTERMODULATION DISTORTION

Intemaodulafion disto~ion is defined as the production of new output signals which are created from the nonlinear combination of two or more input signals. This inte~odulation occurs because of the nonlinearity in the amplification process~ The order of the intermodulation product depends on how many input signals are mixed and which h ~ o n i c s of each of those input signals have mixed. Second-order and third-order intermodulation products are defined as follows. Second-order intermodulafion woducts are

-f~ - A =A U, + A = A f2 -f~ =/4 f~ - A =A, where frequencies f3, f4, and f5 axe the undesirable distortion products that show up at the output. ~ird-order intermodulation products axe

~ + A=f6 ~+¾=~ ~-A=A

6.2. Distortions in TWTAs

211

If fl and f2 are very closely spaced, then the third-order products 2fl - fl are the most difficult to deal with. These spurious signals fail. in the vicinity of f~ and may show up in the receiving passband with sufficient amplitude m cause interference problems. R e fourth-order intermodulation products are

3A

- A =flo

The fifth-order inte~odulation products are

2A

3ji

3f2 -- 2Ji = f t 3 "

These possible intermodulation products were. produced by just two input signals. Figure 6.7 shows a graphical representation of some of the intermodulation products. Finally, Figure 6.8 shows a plot of power output vs power input, not. only for the transmitted signal, but also for the intermodulation products. Notice that as shown in the figure, the slope of the second, order inte.~odu!ation product is two times the slope of the desired signal output. R e slope of the third inte~odula.tion product is three times that of the output signals. Notice also that both of the. inte~.odulation plots intersect the. output level because they are offset from and at. different slopes than the output level. These points where the interseztion occurs are called intercept points.

Signal 1

:. ..............

==: ...................................................

...,,,:.,

3fl " 2f2

,:,.,,:,. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

.

Signal 2

...............................

2fl "f2

fl

f2

2f2 "fl

Figure 6.7 Some intermodulation products from two signals.

3f2- 2fl

212

6. Noise Representations in Transponders and Multiple Access

/•'•t'--

Second-order intercept Power ~ . /~. 7~ . . . . . . . . . . . . . . . . . . . ~ird-order . . . . . ... intercept JL Output: S ~ . .iz/ / g n a l . Output (dB) ... ~" / ' ~ Second-orderIntermod Third-order Intermod

ut Figure 6.8 Input/output power relationship for transmitted and intermodulation products~

6.3

Multiple Access in Satellites

Multiple access in satellite communications means sharing all the satellite's resources. The theoretical and technological procedures encountered in accessing a communication network by multiple stations are referred to as multiple access processes. The methods of transmitting information in a simultaneous manner, including that of point-to-point in earth stations, to share a common resource network are referred to as multiple access techniques. In satellite communications, access techniques allow the satellite to be shared for mulfidestination signal transmission. There are basically three different (often called oahogonal) access techniques. For the technique dealing basically with divisions in time slots, we have time-division multiple access (TDMA). For the technique dealing with divisions in a frequency band, we have frequency-division multiple access (FDMA). When ~ M A is combined with TDMA, we have a time frequency (TF) access scheme, which includes frequency hopping (FH) and time hopping (TH). The third access technique is code-division multiple access (CDMA). CDMA is a TF scheme which makes use of the elements or cells of the time and frequency two-dimensional plane as a result of both the time and frequency divisions. In CDMA the access signal, can be represented by a collection.

6.3. Multiple Access in Satellites

213

of occupied cells in. a time-frequency matrix: of the TF scheme. All three satellite access schemes are shown in Figure 6.9.

6.3.1 FREQUENCY-DIVISION MULTIPLE ACCESS FDMA uses frequency allocation among a set of each stations to share a series of frequencies. The transponder bandwidth is divided into a series of nonoverlapping frequency slots, each of which constitutes an access channel. Because the nonlinearity of the satellite transponder causes intermodulation products, nonlinear signal power transfer, and intelligible crosstalk, the TWTA cannot be operated to full capacity. Therefore, the power efficiency of an FDMA system is decreased because of the n e ~ to decrease the power of the TWTA. Other potential problems in ~ M A are as follows: 1. During • uplink, RF power coordination is needed to prevent the suppression of weaker signal by stronger signals 2. Low flexibility; need to slowly change preassigned, traffic patterns 3. Carrying capacity of traffic is relatively tow

fn+Afn * f fn

...................

............................................................

fl+Afl fl

.........

i i=i

............

" iii~time

(a) FDMA Scheme

f

C4_.4

f _ _

t!

tn

tl+Atl

(b) TDMA Scheme

tn+Atn

I

C23

~. . . . . . . . . . . . . . .

.........

,,,:

._

(c) CDMA Scheme

Fig~Jre 6.9 Different satellite access schemes~

214

6.3.2

6. Noise Representations in ~ansponders and Multiple Access

TIME-DIVISION MULTIPLE ACCESS

The time-division multiple access scheme is an access technique in which, signals from each earth station are. sent into burst (time segment) and in a sequential manner, gaining access to a common satellite transponder within a periodic TDMA frame. Each earth station burst (see Figure. 6.10) occupies a particul~ slot in a basic time frame: without overlapping with bursts from other earth stations. Burst overlapping can be avoided by burst scheduling and network synchronization. The TDMA scheme has the highest degree of satellite efficiency, but requires excellent timing and network control with other eaah stations. TDMA is less susceptible to interference, since it can operate near saturation without producing significant interference. Therefore, in order to saturate TDMA, the transponder must be capable of producing very high power transmissions. In TDMA, at each instant of time there is only one carrier using the transponder, and there is no intermodulation problem. However, one main disadvantage of TDMA is that it requires network timing and synchronization. It is possible to supply vari~ie time slots, but these slots must remain unchanged during operation, limiting the syste m ' s ability to accommodate rapidly changing traffic patterns, because accurate synchronization is needed. TDMA needs long synchronization sequences, which decrease the efficiency of the system.. Furthe~ore, though there are no intermodulation problems, TDMA is still susceptible to nonlinearity problems in the form of intersymbol interference, which needs to be minimized by selective filtering.. One of the main advantages of TDMA over FDMA is that in ~ M A , the earth station must transmit and receive on multiple frequencies to achieve a desired traffic plan. Therefore, there is a large number

N

N

N

N

i II

Figure 6.10 The TDMA scheme at a system level.

N

l

6.3. Multiple Access in Satellites

215

of frequency-selective upconversion and downconversion chain.s. In a TDMA system, the needed selectivity is obtained in. time rather than frequency. In a multiple-beam satellite system, the stations of each beam communicate with other stations in all other beams. If FDMA methods were to be used, the satellite's up-beam must be routed to the down-beams through transponder filter banks and additive combiners which sum the noise from all. up!ink beams, creating an overall low S/N ratio. The use of TDMA allows the use of a satellite switch which selectively connects individual up-beams to individual down-beams, eliminating uplink noise problems. Furthermore, by synchronizing the locations and the duration of individual station traffic bursts and the location and dwell, times of satellite switch beam-m-beam connections, the total traffic among all beams can be accommodated in a simple and optimum manner.

6.3.3

BRIEF OVERVIEW OF TDMA ARCHITECTURE

Each of the traffic bursts shown in Figure 6.10 looks like the frame, shown in Figure. 6.11. Each frame begins with a reference burst. The locations of traffic bursts are referenced to the time of occurrence of the reference burst. Traffic bursts originate from a given emeth station and caxry the traffic from that station to all destinations in a digital transmission format. In Figure 6.11, the start of a. traffic burst from station 1 occurs at time T~ after the reference burst. The traffic burst from station 2 occurs at time ~ after the reference burst, and so on. The position and duration of each traffic burst relative m the reference bm'st are

Reference

'I~~, Reference Burst ,

traffic burst

m

r

•

| .........

" ' T1......" 7

t

i

1

T2

•

!

I

i

I

I

I

I

i

I

I

I '

I I

I

F,q ........................................................ t

Burst

Tn

r

TDMA frame Frame m

Figure 6.11 Example of TDMA frame.

I Frame m+t

216

6. Noise Representations in Transponders and Multiple Access

~

. . . . .

..........

mir, gRooove,4Wo ,,

rvice Channel

.....1 1

....

..........1 i,,0,oo P:lV°cel " .......

Delay Channel

....

"

"

F i b r e 6.12 Architecture of a reference burst.

scheduled to a protocol established by the network. These protocols, however, may change depending on the traffic demand. As previously stated, reference bursts are emitted by a reference station and are the basis for synchronizing all other stations in a network. The architecture of a reference burst is shown in Figure 6.12. h i s reference burst has the information needed by other stations to obtain the exact location of their bursts in the frame. The reference burst is made up of three parts. The first is the "carrier and bit timing recovery," which serves the objective of look5ffg at a received station to the carrier frequency and the bit timing clock of the burst. This is followed by the "unique word." The unique word is used to reference the time of occurrence of a burst, and it also marks the symbol time reference for d~oding information in the traffic p~t of a burst. Following the unique word, the reference burst contains a serv ce channel' and order features which can be used to support operating system protocols and for utility teletype and voice communications among stations. Following all these and last is the control and delay channel, which serves to corr'anunicate information for the control of burst positions to station.s in the network. The traffic burst architecture is shown in Figure 6.13. Traffic bursts are synchronized relative to the reference burst to occupy assigned l~ations in the TDMA frame~ Notice in the figure that the "preamble" of a traffic burst has the same architecture as the reference burst; actually, the c ~ e r and bit t i ~ n g recove~ and unique word are the same as in the reference burst.

Figure 6.13 Tra~c burst arc~tecture.

6.3. Multiple Access in Satellites

6.3,4

217

CODE-DIVISION MULTIPLE ACCESS

Code-division multiple access systems originate fiom a more general class of system called nonorthogonal systems (which also includes spread spectrum and time/frequency hopping). Actually, most nonorthogonal systems have the common property of spread spectrum. In a CDMA system, a message spectrum can be spread by pseudorandom (PR) code sequences to enable users to obtain access through a corp~on signaling channel. There are three basic methods which can produce a spread-spectrum effect: direct frequency, frequency hopping, and time hopping. CDMA is an extension of direct-frequency spread spectrum and frequency-hopping spread spectrum. CDMA systems provide multiple access communications capabilities. In CDMA each user has an individual, distinctive pseudonoise (PN) code. If these codes are uncorrelated with each other, then within the same mobile cell, K independent users can transmit at the same time and in the same radio bandwidth. The receivers decorrelate (or despread) the information and regenerate the desired data stream DM( 0, where m = 1. . . . . k. In Figure 6.14, a CDMA spectral concept of several direct-sequence spread-spectrum carriers is illustrated. If the received power Ps of all these signals is the same, then any one of the desired signals will be interfered by m - 1 equal-power CDMA signals. Therefore, the RF received carfier-to-interference ratio (C/1 (dBm)) = 10 log (l/m), which is a negative number known as "self-interference" caused by the other m - 1 carriers that simultaneously occupy the same bandwidth as the mth desired c ~ e r . The probability of error (Pc) caused by the self-interference of the m simultaneous equal-power received signals is given by Pc = ~ erfc

~

,

(6.8)

where jQ is the carrier frequency, fb is the bit rate, and m is the number of users. It is assumed in this equation that thermal noise is negligible and that the powers received are the same and uncorre!ated codes. ~ e more general error probability for a general CDMA was derived through a theorem of higher moments, and the expression is given by Equation (6.9), where h is the magnitude of desired signal, n is the: zero mean additive white Gaussian noise, z is the interference from m - 1 users, D is the maximum values of the interference variable z, m l = E(z2), m2 = E(z4), and ~ is the variance of n. The bounds can be evaluated once we know the values of h, m m~, m2, and D. The terms m~ and m2 depend on the probability density function of the

6. Noise Representations in Transponders and Multiple Access

218

e Dt(t)

Gl(t)

Dm(t)

Gin(t)

........................

D2(t) ~(t)

]~

[~

D3(t) c~(t)

Din(t)= ruthdatasource(m users) ~(t) = ruthPNcode(m users) l Space

D4(t)

~(t)

DS(0

,.,

tuency Time Figure 6.14 CDMA system with m users sharing ~e same receiver satellite. interference. For PN code sequ~.nces, the density function can be approximated as Gaussian. Notice from the following equation that the error probability is highly sensitive to the number of' users in the CDMA system.

{h + (ml)l/4~

Pe (up~r bound)= effc \ ~ j

+ erfc(~)+

{h-(rnl)"4\]

+ erfc\ ......;..: ~ , ......]... [erfc/h + D)

ert~(h.....--.D't,effc( h__~)]

mt • k,' Y ~ ]

+~

',, m2

~ - m~ .

P~• (lower bound) = effc h +~ r(m2)t!4'] tm2j j| - 2 erfc ~ ~h ' ~ ) + effc h ~~r-~ + 2 erfc ( ) h

m~ (6.9)

6.3. Multiple Access in Satellites

219

A general power-bandwidth relationship of spread spectrum systems is given in terms of signal-to-noise ratio at the receiving detector output after going through a nonlinear transponder of usable bandwidth Bt (Hz): (57N)

=-~

kBm~(~z) /

(6,1,0)

1 ~ (~N)rJ"

Here, (C/N) is the received carrier-to-noise ratio in system bandwidth B t (Hz), Bmc (Hz) is the message bandwidth, and m is the number of simultaneous signals present at the transponder input, all assumed to be of equal power. A comparison of FDMA, TDMA, and CDMA for small satellite network earth stations is given in Table 6.1, where K is the number of accesses. A block diagram of a CDMA transmitter and receiver is shown in Figures 6.t5 and 6,16, We are assuming a modulated binary PSK signal S(O given by m

S~i(t) = ~ ~

Di(0 cos 2 ~fif t,

(6.11)

i=t

Table 6.1 A Comparison of FDMA, TDM~, and CDMA Type of Access

Power Eff:w&ncy BandwMth

FDMA

0.8

10/3K.I~3

Most Uplink power vulnerable coordination. Frequency assignment changes with changing the number of access.

TDMA

0.8--0.95

10/3KI.3

Most Synchronization is Most vuln~able required.Power control not required.

CDM_A

0.8

3//I'~ K Least Need no central ~-~,.C) K - 1 vulnerable control.Frame syncNonization is not essential.

Jamming

Oper~on Feature

Complexity Moderate

Least

220

6. Noise Representations in Transponders and Multiple Access

Z ~_D,(t)Cos2~f ~t:t m

D!(0 i

m

~=~

..

Z

2~./G,(t)D, (t)Cos2~ fret

i~l

O2(t)

O

~(t)

"1 Dm(t) (rate == ~)

GI(I) SpreadingSign~ func~aonGi(t) PN genetator

IF Carrier I A Cos ~ flF t

transmitting antenna

Chip rate, ~ = 1 ~ Upconverter v

m

~ Upconverter

2~-~G,(t)D,(t)Cos2x f o t

i=t

Figure 6.15

m

~F;mP;

...........

Block diagram of a CDMA transmdtter.

•

.............

>

~ q ( t )

•

SK

~ cos~O f~.,,S / ./, l' - ............ -..............

+ '-'

.

i

o

Figure 6.16

Block dia~am of a CDMA receiver,

~i~

t-I,Io

I

6.3. Multiple Access in Satellites

221

where ~f is the can:let frequency, and Di(t) is the unfiltered binary signal (+ 1 or - t ) for each of the m users (bit ratef~, = 1/Tb). The term P~ is the corresponding carrier power. The binary BPSK-modulated CDMA signal is given by

S2i(t) - 7~Gi(t) S~i(t) - ~ ' ~ i=I

Gi(t) Di(t) cos 2 ~ t ,

(6.12)

i=I

where Gi(t) is a pseudonoise (PN) signal (or spreading signal) having a chip rate of f~ = I/Tc. There is a different PN code for each user (i = 1, 2 . . . . . m). Notice that because we are assuming an ideal multiplication in Gi(OSj~(t), this procedure is equivalent to double..sideband suppressed carrier amplitude modulation. The intermediate frequency f f is upconverted by an RF synthesizer to the transmission frequency. In this notationff co~esponds to the IF frequency. Users (i = 1. . . . . . m) within the same cell share the same RaWcarrier frequency and occupy the same bandwidth. The second modulation or spreading is done by a PN coding. The spread spectrum G~(t)S~(t) is upconverted to the @sired RF frequency. At the receiver, the G~(t)Sl~(t) terms are combined with K other independent spread spectrum signals that use the same RF band. The combined received signal is m

R,,~(t) = ~ ' , ~

G~(t) D,(t) cos(2%~)t + 0,~) + N(t),

(6.1.3)

i=l

where m is the number of simultaneous users, 0~ are statistically independent random phases, and N(0 is the noise that stands for both Gaussian noise and interference. The term N(t) can be expressed as N(t) = ~'2¢~1 cos(2 rrf)t + &,

(6.14)

where Pi is the power of the interference signal at the receiver RF input. In these figures, since 0 is a random phase variable and unco~elated, its average value is 1./2 (i.e., cos z 0 = 1/.2.) and the noise term becomes, after the integration filter, <3

cos o, \

¢rftL

/

e, = 4~,r~'

i/l ~
(@15)

Chapter 7

7.0

Satellite Antennas

Introduction

The last exit ~ i n t of a satellite transmission system and the first receiving point of a satellite receiver system is the antenna. We will first address some f u n d ~ e n t a l s of antenna theory and cover some issues concerning noise and cross-coupling (of interference) phenomena among receiver and transmitting antennas. We end this chapter with a good discussion of types of antennas common in. satellites~

7.1

Some Fundamen~ls in Antenna Theory

An isotropic antenna is defined as a hypothetical antenna having the same radiation in all directions (i.e., uniform radiation). It is ~ s u m e d that the power gain of an isotropic antenna is 1.0. The dipole antenna has a power gain of 1.64 or 10 log 1.64 = 2.15 dB above isotropic. A directional antenna is one that radiates or receives electromagnetic waves in some directions better than others. An o~idirectional antenna is defined as one having an essential nondirectional pattern in azimuth, and a directional pattern in elevation. An example of directive pattern is that of the dipole as shown in Figure 7.1. The radiation pattern of antenn~ is described in terms of lobes. Figure 7.2 shows the different lobes of another directional antenna such as a horn. The space su~ounding an antenna is divided into three regions: the far field, the near field, and the reactive near field. The far field is also known ~ the Fraumhofer region and is defined as the field where the angular power distribution is independent of the distance from the antenna, if the antenna is of size L, the far-field region is defined as that with an inner b o u n d ~ given by R = 2L2/~. The near field, also known as the Fresne! region, is that region wherein radiation fields predominate and the angular field power distribution, is very much dependent on. the distance from the antenna. The inner boundaccy is taken to be R -> 0.62 ~,/L3/3, and the outer b o u n d ~ is given by R < 2L2/A. The reactive near field is defined as that. region immediately surrounding the antenna. The

222

7.1. Some Fundamentals in Antenna Theory

~

223

Z

Y

Figure 7.1 Pattern of a dipole antenna.

outer boundary of this region is given by R < 0.62 ~ . In Figure 7.3, we see a diagram of these three regions. T1ne average power radiated (also known as pointing vector) by an antenna is given by P,.~d = ~

Re(E × H) d s. s

Major

Lobe • •

Beamwidth

Figure 7.2 Representation of horn antenna.

(7.1)

224

7. Sa~llite Antennas

Id Figure 7.3 Field regions of a typical, antenna.

For an isotropic point which is radiating with a transmitted power Pt and a power gain of Gt, the power density from the preceding equation reduces to Prad

=

PtGt

4 ~ R 2'

(7.2)

At this moment it is appropriate to. define the concept of power gain of an antenna: Gain = 47r Radiation intensit = 4 ~ U(O, ck). Total input power Pin "

(7.3)

ff we state that the t o : m l radiated power (Prad) of an antenna is related to the total input power (Pin) by the expression Prad = 8t Pin;

where e~ is the total antenna efficiency of the transmitting antenna, Equation (7.3) becomes (7.4)

7.1. Some Fundamentals in Antenna Theory

225

The term 47rU(O,&)/Pra d is defined as the directivity gain of an antenna D(@&). Therefore, G ( O , ~ ) = e t D(O,q~).

(7.5)

In many cases, an approximate formula for the gain of an antenna is given by G(O,&) ~

30,000

~2 ~

,

(7.6)

where 0hi is the half-power beamwidth (degrees) in one plane, and On.2 is the half-power beamwidth (degrees) at a fight angle plane to the other plane. The antenna efficiency et is given by e t = (I -IFI2)eca,

(7.7)

where F = Zin - Z°

z~ + Zo" Zin is the antenna input impedance, Z o is the characteristic impedance of the transmission line, and 8cd iS the conduction and dielectric efficiency. The impedance of an. antenna can best be illustrated by the diagram of Figure 7.4. The generator "v'~has an impedance of Zg = Rg + jXg. The antenna also has an impedance Zout = Rou~ + jXout. Therefore, the figure can also be represented

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

vg

Figure 7.4 Transmitting antenna basics.

226

7. Satellite Antennas

in terms of the electrical network given by Figure 7.5. R~(ant) is the radiation resistance of the antenna, and RL is the loss resistance of the antenna. T,ne total impedance in Figure 7.5 is given by Ztota! = (Rs + Rr(ant) + R L) + j(Xan t + X,).

(7.8)

The current developed by the loop is given by Is(amps) =-(R~+R~ian, i

+ RLi+j

xsi"

(7.9)

The power deliver~ to the antenna for radiation is given by

R....,) = ~

Prad(watts) = ~ Is

[ .......................................................................... ] L(R,~°n~)+ RL + R~)2 + (xo., + x02]"

(7.10)

The power dissipated by heat is given by Praa(Watts) =

1

2

RL

Vs(~

is

Rs

Ran t ,',!b ' = Rr(ant) + RL

Xant [~] ×s

Figure 7.5 Electricalnetwork representation of an antenna.

(7.1!)

7.1. Some Fundamentals in Antenna Theory

227

The maximum power delivered, occurs when Rr(ant) + R L = R s

Brad = ~

and Xant = - X s and Rraa, PL become [ .........Rr(~:!!~) ... .........2] ...

(7.12)

PL = 1VSI28 (R;ii21~'~+ . ..~ , R....) i ~"

(7.13).

(Rr(ant). + R L) j

The power delivered by the source is given by Ps(watts) = Pr(an0 + PL ""~

"

IV~]2[ Rr(~n~:' + R~,,] 8 [(Rr(a,.o + RL)2J"

=

(7.,14)

The effective area of receiving antenna is the ratio of power delivered to the load to the incident power density: Aeff = ~ d

Rr(an~ + RL

(7.15)

In a more general way, the maximum effective area of the receiving antenna can be related to the directivity: Aerr(max) = e ~ D(0, ~b).

(7.16)

~ e Friis formulation relates the power received to the power transmitted between two antennas separateA by the distance R > 2L21~ (D is the largest dimension of either transmitting or receiving antenna). ~ t us consider Figure 7.6.

transmitting antenna

i- i-

C. ~ ot, eL) J

...,, ~qll.,

.

.

i .

.i

~,~

.

.

",,

.

",~

"-..

receiving antenna •

....

(~

.,_OrL -

R ,....,. ~JJ. .....................

Pt, C~, ecdt, Dt, Ft

Pr, Gr, ecdr,

Figure 7.6 ~ e Friis representation for transmitting and receiving antennas.

228

7. Satellite Antennas

For a transmiuing antenna the transmiUed power in the direction of (0t, q~t) is given by Pt(0t,4) = e t

PtD,( Ot,4) 4.rrR2

.

(7.17)

The effective apemare of the receiving antenna is given by A r = a:r Dr(0r,q~r)

.

The .amount of power (watts) collected by the receiving antenna is given by Pr = er. Dr(0r, q~r)

Pt(0t,(;b0

A2 Pr - e~ et D,:(0~,,;b,:) (4"')2 R 2 Pt(0t, 4)Dr( 0t, 4 ) ,

(7.18)

which can also be represented as the Friis trans~ssion equation,

Pr ~-

~cdt 8cdr( 1 -- rt2)( 1 -

rr)~-~-~ Dt( Ot,bt)Dr( Or,~r)Pt.

(7.19)

ff the antennas are matched for maximum direction radiation and reception,

PtGrGtA 2

Pr = ~ "

(7.20)

~ e mrm A can be expressed as A (m) = 300/f (MHz). Substituting back,

Pr-t"

P GrG (300/f~ :z •

t\ 4

[ 569,93 6 ]

(7.2i)

Pr = Ptara, Lf2~nz-~im)2 j. The term. Pr/Pt is known ~ the propagation loss and, in decibels, can be expressed as

fit (dB) = Propagation loss = 10 log G t + 10 tog G r - 20 log f (MHz) - 20 tog R(m) + 27.558.

(7.22)

7.2. Antenna Factor and Electric Fields

229

The electric field in volts per meter and power density Pd (W/m) at any point in the far field in free space are related by Pa =

E 2 (V/m) t207r "

(7.23)

Combining Equations (7.23) and (7.2!), we have E : (V/m)

or

E (V/m) =

R (m)

"

(7.24)

Substituting this last ~uation into Equation (7.21) for received power Pr, we obtain. Pr =

E 2 (V/m) • Gr' (18.9977)

f2 (MHz)

"

(7.25)

The electric field is the field at both the receiving and transmitting antennas. In terms of received power, the preceding expression can also be started as

Pr e (V/m) = y (MHz) ~/'G ;: i~8~i~9.9~.

(7.26)

Other equations of interest are E 2 (V/m)R 2 (m) 30Gt Gt =

E 2 (V/m)R 2 (m) 30Pt

(7.27)

Pr = Pt" log-I (Path loss/10).

7.2

Antenna Factor and Electric Fields

The antenna factor in decibels is simply a measure of how many volts an antenna will output when placed in the presence of an electric field. The antenna factor varies with frequency and is unique for every type of antenna. However, the most important use of the antenna factor is to calculate the electric field by measuring the voltage at Zi of the receiver. Therefore, the antenna factor is the

230

7. Satellite Antennas

"X~~ E(Wm)

Ir , , I,,

Cable Loss

r-I

I

Pr

n

I System

Figure 7.7 Receiving antenna representation for antenna factor calculations. loss created by the antenna in convening the electric field (E) to a voltage as shown in Figure 7.7. The antenna factor AF (dB) when the receiving antenna gain Gr is given in terms of dBi gain is ~ r (dB) = 20 log f (MHz) - 29.78 - Gr (dBi).

(7.28)

The antevma factor (numeric) is given by AFr (numeric) = (9.76 / A ) V ~ r . For a transmission antenna, the antenna factor is givep by -/x~ t (dB) = 10 log G~ - 20 log R - I0 log Zin + 14.77.

(7.29)

The electric field at the receiving input of the system of Figure 7.7 is given by

Ev/m (at antenna)

Ev/m(at Zin) - log_iiAFt_ (Cable loss (dB))/20]'

(7.30)

where E (v/m) at antenna is given by ~uation (7.26). Also, Ev/~ (at antenna) Ewm(at Zm) - log= ll~aA~r - (Cable loss (dB))/20t'

7.3

(7.31)

Antenna Interference Coupling

Let us consider the parasitic scenario of Figure 7.8, Vt is the driving source voltage of transmitting antenna, V2 is the induced voltage due to parasitic coupling

7.3. Antenna Interference Coupling

~Z12

Receiving

f

231

Parasitic Coupling

.Z-_/ Receiving Antenna

Transmi~ing Antenna

#3 .

.

.

.

.

.

.

#1

#2

.

4-

Za3 ~ . . ,

..

'4>-

zl~

Za2

Figure 7.8 Antenna interference coupling model. from. transmitting antenna, Y~ is the input admittance of antenna #i, Y22.is the input admittance of antenna #2, and Y~2 is the mutual admittance of antenna #2 due to coupling from antenna #1. The preceding system can be represented as a two-port network (see Figure 7.9) using an admittance matrix. Y (Y = I/Z): V ! = Y ~ I 1 + Y1212

V2 = g21ii + Y2212, where = input admittance at port #1 with port #2 short-circuited Y12 = llN2!v!=o = mutual admittance at port #1 due to a voltage at port #2 (with port #1 short-circuited) Y21 = I2/VlIv2=o = mutual admittance at po~ #2 due to a voltage at port #1 (with po~ #2 sho~-circuited) Y22 = 12/V2tvl =o = input admittance at, port #2 with. port #1 short-circuited. YII = ll/Vl[v2=o

7, SatelliteAntennas

232

I1 +

~,.-

Vl .iLl ............

::ZII

i

12 .............

..............

½

[Y] i i-~.~iz~

...........

+

•

22222= . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

i i i . i . z

.....

:~.~z.,;T;2;.,72,77.~

......

i

iii..,

i

ii.

7

Figure 7.9 Two-port network representation of antenna coupling.

Coupling: between antennas is often a parameter of interest, especially when a receiving system must be protected from a nearby transmitter. Maximum power transfer between antennas occur when the source impedance and receiver load imwdance are conjugate-matched to their antennas. Determination of this condition is complicated by the antenna interaction. The coupling problem can be solved in closed form by the Linville method [6], a technique used in RF amplifier design. The first step is to determine the two-po~ admittance parameters for the coupled antennas by exciting each antenna with the other short-circuited and then computing the self and mutual admittances from the measured currents. The maximum coupling is then Cma x =

Couplingmax = ! [1 -- (1 -- L 2 ) l a ] ,

(7.32)

where g

2 Re(Y]~). Re(Y22) - Re(YI2Y2t)" The matched load admittance on antenna 2 for maximum coupling is

l-p

YL . . . . i = : ; p +

I

]

Re(Y22)-~ Y22,

(7.33)

where = Cma_x_(Yt2Y21 )* P

IYlaY21[

'

where * stands for complex conjugate and the corresponding input admittance of antenna 1 is ~n-

Y2~Y12 Y11- YL + Y22"

(7.34)

7.4. Satellite Antenna Systems

7.4

233

Satellite Antenna Systems

Satellite antennas can provide communication paths with other satellites ~ well as earth stations. It can be shown that the average coverage area (Acov) of a satellite (see Figure 7.t0) is given by Acov ..~ Pt Aes(C/N) .... 1 =

B (Hz) To L~ '

(7.35)

where Pt is the satellite transmitting power, B (Hz) is the bandwidth in Hz, (CIN) is the career-to-noise power ratio, Aes is the earth station antenna effective area, Te is the system, teml~rature, and L i includes all losses. Notice that incr.e.asing the coverage area requires a reduction in bandwidth, which is not always desirable. If we reduce the coverage area (e.g., using a multibeam antenna), we can increase bandwidth, which is very desirffble. A typical satellite communication antenna consists of three main elements: anmnna structure, feed sysmm, and beam-forming network. The antenna structure

I I I !

\ \

\

\

\

Figure 7.10

Area Covered (Acov).

Satellite coverage area.

234

7. Satellite Antennas

can be an array, a lens antenna, or a reflector antenna. The feed system is strategically located near a focal point for maximum direction of energy. The beam-forming network distributes the energy to. the proper output port and conrains power dividers and switches. In a communication system using TDMA, high scanning rates need to be achieved in order to provide enough generated beams. To accomplish this electronically, control switching must be used. A fixed beam can be a pencil beam or a shaped beam. Shaped beams are used m improve antenna gain over a specified area to reduce interference outside the intended coverage area. The purpose is to obtain beams that are shaped, for maximum antenna gain with minimum side lobes. The three basic antennas,: as previously stated, are the phased array, the lens, and the reflector. Table 7.1 shows the advantages and disadvantages of such antennas in satellite c o ~ u n i c a t i o n s . We will concentrate in. this section on reflector antennas, which are the most widely used in satellite communications because of their simplicity and low cost. "We first, however, briefly introduce the phase array and lens antennas.

7.4,1

PHASE A R P d Y ANTENNAS

Several antenna elements radiate in phase coherence. Among ~ e elements we can find axe horns, dipoles, helices, and spiral antennas. Phase arrays can be used for cases such as a fixed beam, either single or multiplexed; electronically

Table 7.1 Advantages and Disadvantages of Three Basic Satellite Antennas Antenna 7Xype

Advantages of Antenna

Disadvantages of Antenna

Phase array

Distribution power amplification at the elementary radiation levels Increased reliability No aperture blockage

Complex Heavy Higher beam-forming losses

Lens

No feed blockage Better scanning performance

Heavy in low-frequency applications Aperture mismatch

Reflector

Simple Lightweight Design maturity

Offset to avoid feed bl~kage Poor scanning performance

7.4. Satellite Antenna Systems

235

steerable scanning beams; or a feed array of a lens or reflector antenna system. The number of elements is inversely pro..portional to the size of the elements. Element size should be kept less than a wavelength, requiring a large size of radiating elements. The arrangement of elements can be a square grid, rectangular grid, triangular grid, or even random arrangement. The element types determine the array gain that we can achieve. Possible candidates are the waveguide, ridge waveguide, helix, and slot a~ay. 7.4.2

THE LENS ANTENNA

The lens and reflector are used as collimating elements in a high-gain antenna system. The lens antenna requires two surfaces to collimate the beam. The surfaces in general have the single form of a plane, spheroid or parabolic. The most widely used lenses in spacecraft applications are waveguide, TEM, and dielectric. A waveguide lens is limited to narrow frequency band operation. A TEM lens is heavier, but has wider bandwidth, and the dielectric lens is the heaviest. The dimensions of the lens are described in terms of the diameter and the ratio of focal length over diameter. The diameter of. the lens is determined by the gain and beamwidt_h r~uirements; the longer the focal length, the better the scan ~rformance. The surfaces of conventional lenses are planar, spherical, or paraboloidal. A s h a r d surface is widely used to obtain the better performance that, with a conventional lens design, would be difficult to obtain. 7.4.3

THE REFLECTOR ANTENNA

The reflector antenna is the most popular in spacecraft antenna systems because of its structural simplicity and light weight. It is also a matured design. The main disadvantage is that the reflector needs m be offset to avoid blockage of the feed point. This offset eliminates the rotational symmetry of the optical aperture, and the scan range is limited to a few bandwidths. A reflector antenna can be made of several reflez.tors, whose surface can be parabolic, hyperbolic, ellipsoid, or spheroid. The most popular reflector antenna is the parabolic. The general geometry of the parabolic receptor is shown in Ngure 7. i i a. For an offset-fed parabolic reflector, the offset axis is in the direction of 0o, where 0u and 0L are the angles subtended at the local point by the u p ~ r and lower edges of the reflector: t

0o = 7(0u + 0L). The focal surface is the plane which is pe~endicular to the offset axis.

(7.36)

236

7. Satellite Antennas

The basic polarization generated by the feed point is that of a circularly polarized beam. When. such is the case, no cross-polarization will appe~ in the radiated electric field from the reflector. Reflector antennas are not good for continuous beam scanning, as beam degradation does occur. Degradations include beam-shape distortion, gain, loss, beamwidth, and higher side lobes. The size of a reflector depends on the gain and beamwidth r~uirement and whether we have a single-beam or multi.~am system (larger reflector required). R e most popular type of reflector is the single offset parabolic reflector as shown in Figure 7.11 a. A spherical reflector, however, is fi-ee of coma and astigmatism. A design somewhere in between is probably most desirable. The reflector surface can either be solid for circul~ly polarized waves or gfidded for linearly polarized waves. Because of the very limited, scan perfo~ance of single-reflector antennas, a dual-reflector antenna has one more degree of freedom and is capable of reducing spherical phase aberration for better scan performance. The most popular dual-reflector antennas are the Cassegrain and Gregorian antennas. Nobabty one of the most important parameters in a reflector antenna is the ratio of the focal length to antenna aperture size (fM), where the main aperture of the largest circular reflector is derived. The larger the f/d ratio, the better the perfo~ance for using a scanning beam. However, this implies a larger antenna system, which may not be cost effective. Approximation f o ~ u l a s have been used [7] for designing an offset parabolic reflector with a circular feed-array configuration. For a given side lobe (SL) level of a single-pattern, half-power beamwidth 2Oo, maximum scan angle 0~ with allowable GL dB scan loss, spacing d between adjacent elements, and offset distance h, we can determine the aperture taper and efficiency reflector diameter, f ~ a l length, beam deviation. factor, and element number. The formulas are given in Figures 7.1 l a-c and Table 7.2, where 0 < ~ < 0.85 I~
=

tan-

I(D/4~

D/4F c = length of correlation interval A = wavelength

7.4. Satellite Antenna Systems

237

ilLz

d2

h

d

L..

.........................

~

I

Y

f Figure 7.11a

General geomet~ of a parabolic reflector.

D = antenna diameter 0 = pattern angle fi2 = phase error variance Az = surface tolerance F or f = focal length. Often we can get reflector surface errors due to distortions in a reflector surface. Two sources of distortions are random surface errors produced in the manufacturing process, and deterministic surface errors due to thermal distortions of reflector surface. Tolerance. t h e o ~ [8.] shows that the effect of random surface distortions results in a reduction of peak gain and increase in side-lobe levels. ~ e pe~urbed radiation field due to the random surface distortion is given by P~ - (27rc / A)2~2e .....(m./A sin. ~2, where 8 2 = (47ra 2 AZIA)2 a

tan ......t(dl4f)

(d/4f)

(7.37)

1.0

0~8

q-PERCENT 80 90

70

,..

_

100

\~q

' c~'-'-k .... "-,,_

,,, . . . . . . . . . . . . . ..............

1,6

K(D~/2) sin e~ 1,8 1,9

i~7

.............................................. ' .......

............" Z

,

2,0

.....

L

0.6

'

",.'\

a ..............

0.4

7 7

. . . . . . . . . .

i

l/i

~I~ X .......... SLI/ i

02

VI~,

1........ ',,

o

-25

-20

~,i ......... ' , / 7

-t 5 -t0 ET-DECIBELS

-5

0

1,6

-

..

18

20

22 24 SL,I,-DECIBELS

K(D!/2 ) sin 0 t 1~8 1,9 2,0

1,7

2,1

26

2.2

1.0 0,8 . . . . . . . . . . . . .

~

......5'i

-

0,6 ........

Z

\

0.2

sk,Z \,7

.--,~;;,.,~,,,..........

*i

0 I:1

i

.............

I

/ .......................................................... ,....

]

,

!

20

22

24 26 28 SL-DECIBELS

30

32

"

0,8 ......... -1...... !.

0,6 . . . .

.

I

ET-DECIBELS 1.0

.

Z

......

..............

0 18

-5

............. i

.

it

...... / ................... :=

v.

7 : Z

0,4

........... ___. - - ........ J

~

.......... ......' ' - 4

20,00 ~'~..................................

. . . . . . . .

13,98

, -------~ t0,46 \-

"

"\%

0,~ ....... . . . "~, ........ . . . . . .[ . .

7. 96

W

6.02

J O

~\-- 4,44 l 3.t0

0,2----

"

I i,94 0 ~,92

¢

Figure 7,11b

50

60

70 80 n-PERCENT

90

100

0

Universal curves for designing an offset-fed parabolic reflector antenna. (a) Relation between A and reflector performance parameters when feed element is of type A. (b) Relation between A and reflector performance parameters when feed element is of type E. (c) Aperture efficiency versus edge taper when feed element is of type B, 238

7.4. Satellite Antenna Systems

/

60

239

50, L

40

/

i1 iiiiiiiiiiii¸IIIIII 1

..............................ii

20

,

i0

__ZZ ........................

0

Figure 7.iie

Z4.4

! I

= : = ~ : : - -

.............

1

....................................................

.......

"

L L .............................

-t.00

-0.75

,,~,,,J,.............................

-0,50

x/F

-0.25

0

Universal chart of parabola.

FEED SYSTEMS

The design of the feed array differs significantly between scannings-beam and fixed-beam antennas. In a scanning-beam antenna, system, design requirements are gain, gain tipple, side lobe, and crossopolarization levels. The gain and sidelobe levels are controlled by the optics in the feed illumination taper. To improve gain. we can use a larger fe~d element, but this can result in a wider feed separation, lowering the crossover point of the two adjacent scanning beams and further increasing the chances of gain ripple. To maintain high gain and reduce tipple, an overlapping cluster feed is used in which a single beam is generated by several feed elements. Continuous scanning is the other approach used to maintain high gain with the help of phase shifters and power dividers controlling the beam position. 7.4.4.1

B e a m - F o x i n g Networks

Beam-forming networks are n-to-m po~ networks with the objective of intercon~ necting the n input ports to the. individual m po~s with. required amplitude and phases. Beam-forming networks can be divided into scanned and fixed beamf o x i n g networks. Scanned beam-forming networks use either variable phase shifters or interconnecting switches to select a beam. Other methods include variable amplitude networks, scanning with both variable amplitude and phase

240

7. Satellite Antennas Table 7.2

Approximate Design Formulas for an Offset Parabolic Reflector

.................................................. : ........... ==. ..............

: .::.............................................................................. - ....... = ............................................ : = = . , .:.,

Design Parameters

Formulas

, po ore

ao = a~ = a2= a3 =

t ~ O

Aperture(r/) efficiency

Reflector diameter (Dr~ A)

( ~A "fl,~3

I

)

X

3

rr sin 05 ,;.~o

1@%

ynN;

Focal length ( ~

7r(sin 0y'sin O!)D 190C cos-t[l - (GU5)]

Beam deviation factor

7"(1 - 0.72e - 3`2(F/rD'))

.

::::

.

.

.

.

.

.

.

Type A (Feed)

.

.

.

.

.

:

.=-:--:-==:

Type B (Feed)

-26.55 35.17 -15.59 2.37

ao = a~ = a2= a3 =

-8,87 9.32 -3.0 0.32

Numerical fiI = -0.026 fi2 = 0.039 f13 = -0,263 Yo = Yl = Y2 = Y3 = C

=

~1~ ~

1.609 0245 -0.259 0.396 t

-

e

c u r v e s

y0 = y~ = Y2 = Y3 =

1.61 0.57 -1.43 1.47

- ' ' O ~ t 2 V ~

cos 0o + c o s ( 0 - O0 i + cos(Oo- oL) .......... . . . . = . . . ..........................

,.<%

...... 1

tan Number of feed elements (N)

Oo(~)[1. =

- I(~214', F] J

Nearest integer of (~/202)

networks. An example of a nonoverlapping switch beam-forming network is shown, in Figure 7.12. A more complex switch matrix which allows choosing one out of n available beams formed by an array is shown in Figure 7.13. A fixed beam-forming network is one in which a fixed beam is formed by feeding one or more feed elements, The number of feed elements depends on the shape of the beam, gain, and/or side-lobe levels. Constrained b e ~ - f o r m i n g networks use transmission lines to transfer energy from input ports to output poas. Unconstrained beam,forming networks use free space ~ a transmission medium. B e a m - f o x i n g networks have the advan~ge of accurate control of

7.4. Satellite

I

2

Antenna

Systems

241

n

iii..... Figure 7.12 Nonoverlapping switch beam network. amplitude and phase. The advantage of an unconstrained beam-forming network is that the size is at. least half that of the overall antenna system, 7.4.4.2

Multibeam Antenna System

The multibeam antenna system is the most widely used in satellites. It is made up of focusing optics illuminated by an array of feed elements. Each of the feed

--

'

i ~~ ? i I

•

ob?_o

~o ° ~-..........°...". ...............

i 1

-°-i

,':

t t 1

Figure 7.13 Switch matrix for beam-forming network.

242

7. Satellite Antennas

elements in the system is responsible for illuminating a single optical aperture and generates a resultant beam. Any shaped beam can be obtained from a series of component beams using the principle of supe~osition. An example of a multibeam system is shown in Figure 7.14. ~ e advantages of such a system are generation of a multibeam pattern from a single optical aperture, pattern, shaping and pattern weighting for the radiation pattern, and steeper pattern roll-off resulting in a higher spatial isolation, within the communications subsystem. Several possible optical, configurations for a multibeam antenna system are shown in Figure 7.15. The power gain of a multibeam antenna system from. using feed elements, each producing a field E,@, 0, ~), is given: by 2

4~R 2 ~ p2(0, ~b) =

~Ei(r,O,~)

i= I 377 ~ l ~ , t 2

(7.38.) '

where Vi are the optical coefficients to the feed elements. The terms ~(r,0,~b) generated by a feed element can be obtained from the induced current present in the driving elements. Finally, Figure 7.16 shows a design procedure for the design, of a reflector antenna.

7.5

Unfurlable Antennas for Use in Mobile Communications

Over the past 20 years, unfuflable satellite reflector antennas have been designed for use in satellite communications. These cover a diverse series of applications

Feed n

zl

.................

)

F d3

N\,,

Feed 2

Feed 1 Individual Beams F i b r e 7.14

Multibeam beam-forming network system..

7.5. Unfurlab!e Antennas for Use in Mobile Communications

m ~ •

I

243

\\ " \\

,/i

"

BeamAntennas

k

""

\\

•

.............. =~ Antennas~ Beam

k

-

, ~

-

\ -~

Figure 7.15 Some configurations of a mulfibeam antenna system.

such as data transfer, land mobile communications, and direct broadcasting. Antenna diameters range from 3 to 150 m, and wavelengths from 1 to 4.0 cm. For example, the Ha~is double-mesh 4.8-meter unfurlable antennas have been flown on the NASA Tracking and Data Relay Samllite, operating at S- and Kband. Typically, the applications require offset configurations with multiple beams. with low side-lo~ and/or low cross-polarization requirements to allow for m.ulti~ ple reuse of the same frequencies. For these applications, the choice of reflector diameter is constrained by the angular sep~ation of frequency reuse zones and, for a given diameter and frequency,, the number of gores or facets and their tolerances derived from acceptable side-lobe level degradation. Typical antenna requirements for some of these satellite missions are outlined in Table 7.3. In. addition to meeting requirements derived from RF specifications, unfurlable antennas must survive the severe, launch and space environment conditions and maintain their surface profile over the lifetime of the satellite. For this reason the bulk of the design, manufacturing, and testing problems are in the structural and thermal areas.

~4

7. Satellite Antennas

Antenna Requirements --Frequency band --Polarization --Coverage area --G(dB) --Side lobe levels _~_~____~*

t~

•

-

'~

.'

"

.....................

Transform the coverage area into antenna

Find ape~ure size and focal length

Determine feed array configuration number ~_.__number of feed e ~ , e n t s

. . . . . . . . . . . . . . .

Develop beam pattern _

.......,...

,~,.=~,o,..,::

~,,,,~,;,,

. , : ...................................................................................................................................... ~:

Perform antenna pattern synthesis No

No/Req

"-,. "es

Figure 7.16 Design procedure for reflector antennas.

END

7.5. Unfurlab!e Antennas for Use in Mobile Communications Table L3 ...............................................................................................

245

Unfurlable Satellite Antenna Requirements

. : -

: . ================================================.=== . .

.

.

.

.

.

.

.

.

.

.

....=:==-.

.................................................

=,::::::::=::.,,,,,,,,,,,,,,,,,,,,,,,,::,,

: ....................

Antenna

Mission

Frequency

Diameters

Mobile communications UHF

820./870 MHz

5 to 30 m

Offset

Replaceable by 1.6 GHz

Sound broadcasting

I..5 GHz

6 to 12 m

Offset

Or 20 to 60 MHz

Mobile communication L-band

1.55/1.65 GHz

6 to 12 m

Offset

Regional coverage

Fixed communications

4/6 GHz

4 to 8 m

Offset

Intelsat

DirectTV broadcast

t.2/17 GHz

3m

Offset

For low package volume

Data relay

15 or 27 GHz

4 to 8 m

Mission dependent

Possibly 2.3 GHz added

Fixed communication Ka-band

20/30 GHz

3 to 6 m

Offset

Muttibearr~hopping

7.5.1

Type

Comments

EXAMPLE AND ANALYSIS OF UNFURIABLE ANTENNAS

The principle and the main constituents of the I~ckheed-Martin wrap-rib antenna are shown in Figure 7.17. The reflector consists of a central hub and deployment mechanism, lenticular cross-section parabolic fibs made of C ~ R , and a goldplated lightweight molybdenum wire mesh. The feed in an offset configuration is typically supported by a deployable mast. A possible application for this technology is for mobile communications where an offset 20-meter, 20-rib reflector with. a focal length to diameter ratio of 1 and 7-mm surface accuracy is considered. The radial-fib antenna (Hams, Figure 7.18) uses pivoting tubular ribs supporting the gold molybdenum mesh. It is flown presently on NASA TDRS satellites for operation, at S-band and K-band. In a simplified version, the mesh surface is profiled by the parabolic fibs. Surface accuracy can be improved by in~oducing an auxiliary system of tensioning quartz wires and catenaries, as is the case for the TDRSS.

246

7. Satellite Antennas

" , . ~,:. ~,: ~'++,. %X. ~,. "~,Nt.2~.~. +t" a~

.~= x ,,

•

',-':.'%+,,

•, ~ + .

.

.

.

.

.

~, ....

,, . ~ .

.

.

.

•

.

.

.

.

.

.

.

~

~ ~ . ~

~.

.... ..... . "

,

•

,+~

:[.e'...e ;/:..s,.'...,P',,, +

•

.~,,. ,,,, ~

,~,

~ ~+P.~I~;~~~ , ~ , ' , ~ ~ c # ~ # M ~ ~

,

,,

, : . . , "~'.. ~.

,~., , ~

W.

.

~,~., . . ~

. . . .

~

,

~ - ~ ~ . , S

. . . .

,,

,., ,..<.

.

...,

.......

",-~ ...,,~+ ;~J:+~.:ii,', V~it :.i+~+'V~2~%:2",,,,,2?..;,."

.

Figure 7.17 Wrap-fib antenna configuration.

There are two areas specific m unfurlable antennas requiring special consideration during the RF design/analysis phase: the reflecting mesh. and the gored: reflecting surface. ~+e reflecting mesh has to be modeled to evaluate its reflection and transmission dyadic, taking into account mesh pattern, wire diameter, contact imperfections, and ohmic losses. Two models are common: the wire-grid model and the strip+aperture model. In the wire-grid mesh model, the mesh surface is assumed to be locally planar, equivalent m a rec~ngular wire+grid mesh. Reflection and transmission of square wire grids were first evaluated by Kontorovitch [9], who derived transmission and reflection dyadic coefficients in closed form. using the method of average boundary conditions. This method was extended to rectangular lattices by Astrakhan [ 10]. The Kontorovitch/Astrakhan reflection coefficients were first used for unfudable antenna mesh surfaces. For rectangular tricot meshes, good agreement was found between computed, results and measurements performed on many mesh samples at normal and oblique incidences with various polarization orientations. For meshes with no clear similarity to a rectangular grid, the four parameters of the equivalent wire grid can be derived from a few measurements on a planar mesh sample. Dumont [I 1] has made an extensive: study of mesh reflective properties, extending the Kontorovitch/Astrakhan theory for wire grids. In addi-

7.5. Unfurlable Antennas for Use in Mobile Communications

"

Figure 7.18

j

247

.afq ~'

Radial-rib antenna configuration.

tion to the four parameters just mentioned, he includes wire conductivity and contact impedance and covers nonorthogonal wire meshes. Complete expressions for reflection c~fficients are given in [12] and the influence of the various parameters is studied. The strip-aperture model in unfurlable antenna mesh was introduced in [13]. Figure 7.19 shows the corresponding lattice and unit cell with the six model parameters, not four as in the case of the wire-grid model. This increase from four to six parameters not only allows better modeling of certain shapes of mesh unit cells, but also allows treatment of nonreztangul~ lattices. The field in the aperture is modeled using a two-mode expansion, sufficiently accurate because of the small size of the mesh cells relative to wavelength. ~ e field outside is expanded in Floquet modes, and continuity conditions are used to solve for the unknown coefficients.. Comp~ison between equivalent wire and strip grid models gives excellent agreement for the proper wire-diameter-to-strip-width ratio.

248

7. Satellite Antennas

Ym 0

0 Xm (a) Lattice

a

y~

Aperture

I

........... ..........

x'

,,7o_

~ConOuoor

...............................................................

(b) A Unit Cell Figure 7.19 Strip-aperture mesh model and antenna analysis method.

The main conclusions from the preceding studies are (I) a wire grid with nonorthogonaI wires can be modeled exactly by an equivalent rectangular grid; (2) increasing the wire grid opening increases cross-polmzation and decreases the copolar reflecfivity for both square and rectangular lattices; (3) departure from a square lattice quickly induces on-~is cross-polarization increases for reflectors with a single mesh orientation and degrades the copolar refle~dvity; (4) refle~tivi~ and polmzation purity increase with the grid wire diameter; (5) poor mesh conductivity increases losses, but only very slightly degrades

7.6. Multibeam Frequency Reuse in Mobile Communications polarization purity; (6) increase in contact impedance impact, but can lead to resonances for discrete values; mesh orientation of different reflector gores breaks down energy in a number of smaller lobes; and (8) the main on cross-polmzation.

7.6

~9

normally has little (7) change of the the cross-polarized impact of mesh is

Multibeam Frequency Reuse in Mobile Communications

In mobile satellite systems, serving the number of channels required depends on a number of factors: (1) the amount of traffic, (2) the nature of traffic distribution for each region of service, (.3) the isolation between users of the same frequency, and (4)the antenna configuration, that provides beam isolation. Spot beam channels are a few tens of kilohertz wide and cover all em'th regions in. a beam configuration. For optimum frequency reuse and least differences in powersharing among all transponders, a combination of spot and shaped beams was found to be e~icient,= An example is shown in Figure 7,20.

CONFIGURATION 1

CONFIGURATION 3

CONFIGURATION 2

CONFIGURATION 4

Figure 7.20 Spot beam antenna configuration.

250

7. Satellite Antennas

: :;:::: :.

;:::! :

:~::::?

21 " '~ .t..~!~iiIii~I;Ii i~-t? .......

i ~i

l]

......... ,o) i

F i b r e 7.21 Global coverage with three spot beam satellites.

Figure 7.21 shows the layout of the three ocean coverages, indicating overlapping regions between the different regions. ~ e total number of user channels (Nu) is given by N u = FR × NBw,

(7.39)

where FR is the frequency reuse factor and N~w is the n u m ~ r of bandwidth channels avNlable.

Chapter. 8

Space Environment and Interference

8.0 Introduction The material in this chapter is based on the work of outstanding researchers in the field of space physics as outlined in Refs. [i4-37]. The region of space dominated by the plasma surrounding the earth is called the magnetosphere. A popular visualization of the magnetosphere is shown in Figure 8.1.

Interplanetary Magnetic Field,~

Magnetopause i Tail Curre~ Cusp

M,AGNEIICALLY PABTtCLES

Field All! Current

Ptasma

" ~ . . . - . . ~ Convecting Plasma Magnetopause Current i Ring Current , /

Figure 8.1 Magnetosphere geometry. 251

Neutral ~ Current

252

8. Space Environment and Interference

The plasma in the magnetosphere can be divided into three regions: (1) the plasmasphere, containing cool (less than 1.oeV) plasma consisting of electrons, ions, and oxygen ions; (2) the plasma sheet, containing warm (5-keV) plasma, and (3) the radiation belts, with plasma energies up to several MeV. The auroral oval is that region where the plasma from the plasma, sheet and radiation belts extends to low altitudes. The space environments for which spacecraft charging is a concern are encountared in geosynchronous and low-altitude, polar orbits. In either of these orbital regimes, spacecraft charging msults when a spacecraft encounters a plasma population associated with a geomagnetic, s u b s t o ~ - - a warm plasma with particle energies in. the t- to 50-keV range. The spatial and temporal variations of the plasma environment in. either the geosync~onous or polar regime are quite complex,, and a large body of work has been devoted to characterization of those variations.

8.1

Geosynchronous Environment

Table 8.1 shows the environmental parameters at geosynchronous altitudes during a quiet wriod. Spacecraft charging at geosynchronous orbit generally ~ c u r s when the spacecraft is enveloped in. the "plasma cloud" injected near local midnight during a magnetospheric substorm. This plasma cloud may be characterized by a low density (1-10 particles cm .....3) wi~ energies of 1-50 keV, in contrast to the "quiet" plasma conditions of higher density shown in Table 8.1. Since a substorm

Tab~ 8.1

Typical En~ronmental Parameters at G~synchronous Altitudes

Plasma density Plasma thermal energy Debye length Ion species Ram ion current, Ram ion energy Ion roach humor Magnetic field Debris and meteoroids Photoemission current

I0 s m- i 0.1 eV 20 cm H+ 5 × 10-8 Am- ~ 0.05 eV 0.7 10'°3 gauss Little 30 #A m '~z

8.1. Geosynchronous Environment Table 8,2

253

Environmental Parameters During a Typical Substorm :-=-=-=-: -.-=-=-=-. : ..:=-: .............:-: ................................. :., . . . . . . . ::.:;::; :

::;:;;;;. . . . .

;:.,,,,,

. . . . . . . . , ; , . ..........................=-...: : = - =

Charging current intensity Charging cu~ent directionality Charging plasma density Ch~acteristic plasma energy Spectrum Background plasma density Time spacecraft in most disturbed region

............................. : ...............:;;;;jj. . . . :;::::.,

t to 1.0 #A m - : Isotropic I0 ¢' to 107 m ..... 1 to 50 keV Broadly distributed No background plasma 30 rain

typically occurs every few hours, the condition for spacecraft charging at geosynchronous orbit exist quite often. A geosynchronous spacecraft can be immersed in: the substorm cloud for many minutes to hours, causing possible duration of hours for geosynchronous charging events. When the plasma is thin or tenuous, the spac~raft charges more slowly than when the plasma is dense. Table 8.2 shows environmental parameters during a typical substorm. Although the spectrum of a geosynchronous plasma is quite complex, it is usually described in terms of a Maxwetl-Boltzmann distribution, either a Maxwellian or a 2-Maxwellian (two populations each represented by a Maxwellian distribution) for each species (see Figure 8.2).The first four =moments of such a

10 ~° _ _

7

> • L A

109 -

U) /

@4

E

,,-X

,_=

i

\

/

m =

10 8

\

/

"

I¢,

107 :......~..... .. 10

....1 ........................................

l

:"-:1

100

I ...................................

10:3

i

.......................

104

t . . . . . . . . . . . . . . . . . . . ;. LLLL'J.JL. . . . . . . . . . . . . . . . -...|

105

Energy (eV)

Figure 8.2

Spectrum for Maxwellian and 2-Maxwellian distributions.

106

254

8. Space Environment and Interference

distribution then equate to the number density, number flux, energy density, and energy flux, which can be compared to actual observations. This parameterization of the plasma affords a convenient means of describing the average plasma conditions, the standard deviation about that average, and the worst-case plasma conditions. For risk analysis, the geosyncbxonous plasma can be described by a Maxwellian [14hi. Table 8.3 gives a 90th percentile Maxwellian representation of the geosynct,~onous plasma environment. The probability of obse~ing given current densities and temperatures of a given magnitude or larger are given in [14]. The SCATI-IA satellite determined that significant levels of spacecraft charging (greater t h ~ - I ~ V) can occur under the following conditions' • Between t900 and 0900 local time • Any distance between 5.3 R,~ and 7.8 R c (Re is the earth radius) - Any magnetic latitude between + 19 and - 1 9 degrees • Any L-shell value between 5~5 and 8.6 • Any period where K v >- 2+, where Kp is the magnetic activity index Charging outside this region can occur at any time electron fluxes at energies between about 30 and 70 keV exceed 6 × 102 cm -2 sec ......~ sr-~ eV .......~ in a plasma-sheet-ffpe low-energy p ~ i c l e environment. A worst-case event occurred on March 13, 1989, at 0700 UT: the "Great Magnetic Storm," where the magnetosphere was compressed from 10 R e to 6.6 Rc. Satellites in geosynchronous orbit were exposed directly to the solar wind. There were several other storms during the peak of the solar cycle. There is no information on particle fluxes or distribution functions below 2 MeV presently available.

Table 8.3

Severe Geosynchronous Substorm Plasma En,~onments

Electron density Proton density Electron temperature Proton temperature Electron current density Proton cu~ent density

1~12 cm ~~3 0.236 cm ~'-3 12.0 keV 29.5 keV 0~33 nAcm -~ 2.5 pA cm -2

8.2. Auroral Environment

255

When doing a post-anomaly analysis, it is usually desirable to use an environ.mental description that fits the: actual environment. Two-Maxwellian distributions are better approximations to space plasmas than Maxwellian distributions (since they have more free parame!ers). This representation, in most cases, fits the data quite adequately over the energy range of importance to spacecraft chm'ging. It inco~orates the simplicity of the Maxwellian distribution while maintaining a physically reasonable picture of the plasma.

8.2

Auroral Environment

The plasma environment of a spacecraft in low-altitude polar orbit is more complex than that of geosynchronous spacecraft. Table 8.4 shows the environmental paxa.meters in the auroral region. The auroral zones are. characterized by visible auroral displays and intense particle and field variations. The necessary conditions for charging appear to be a thermal plasma density less than 1 . 0 4 c m .... 3 and a high integral electron number flux for energies greater than 14 keV. The environment at auroral latitudes in the ionosphere is different from that of geosynchronous orbit in two major ways, First, there exists a large reservoir of high-density, cold plasma that tends to suppress charging effects by provlding an ample source of neutralizing current. Second, auroral electrons are often observed to undergo field aligned accelerations of several kilovolts. Severe charging environments at auroral, latitudes, are more intense. Since a low-altitude, polarorbiting spacecraft is in the auroral region only part of the time, severe charging

Table 8.4 Enviromental Parameters in the Auroral Re#on During Quiet Tim~ ...................................

.............................

Plasma density Plasma thermal energy Debye length Ion species Ram ion cun'ent Ram ion energy Ion roach number Magnetic field Debris and meteoroids Photoemission cu~ent

=

10t° to I0 t2 m ''3 0.1 eV t cm O +, H +, and others 5 × 108 A m :~2 5 eV 7 0.3 to 0.7 gauss Significant. 30 #A m2

256

8. Space Environment and Interference

events occur more. frequently than on geosynchronous spacecraft. Table 8.5 shows the environmental parameters during an aurora. Using several million high-latitude spectra from the DMSP satellites showed two distinct regions of plasma [ 151. The hot plasma that causes spacecraft charging is in an annular region about the pole. The plasma between the poleward edge of the annular region and the pole is colder. The most severe charging environments for low-altitude, polar-orbiting satellites are associated with wes~ard-traveling surges and inve~e.d-V events. Within the poleward bulge of the westward surge, significant numbers of electrons, extending out to high energies, appem" m be present. This indicates that the bulge region may be a severe charging region. ~ e r e a s the westward-traveling surge is generally localized to the night side near local midnight, extending toward local dusk, t~he inverted Vs. have been reported at all magnetic local times [16, 17]. MuUen and Gussenhoven [ 18] showed that the most severe charging events are associated with strong fluxes with energies greater than 10 keV. in two inverted- V events in January 1983, the satellite was shown to charge significantly, mad the charging level appeared to be directly correlated with flae integal flux of electrons over 10 keV. The severe charging environments appear on the night side of the aurora and can have electron current density values up to 102 nA cm.-2 and characteristic energies of up to 15 keV. The ambient, thermal plasma p~icles with energy under 2 eV also v ~ . At times, severe auroras can be accompanied by low ambient plasma density; It is during these events that the highest spacecraft potentials develop. The charging events, in the auroral region, tend to be more intense than at geosynchronous altitudes, but are of shorter duration, tens of seconds rather than minutes. Charging over 100 V is likely to occur under the following conditions: (1) The plasma density is less than 104 cm -3, and (2) the

Environmen~l Parameters Du~ng Aurora

Table 8.5 •

_.

.......

===============================================================

....

Charging current density Charging current directionality Charging plasma density Ch~acteristic plasma energy Spectrum Background plasma density Time spacecraft in most disturbed region

,:.,: .................

--:

.............

I00 m A m -~ Anisotropy

- -

~ ............

t0 6 tO 107 m -3 I to 100 keV Accelerated distribution lOs to I 0 9 m - 3 Under 1 rain

8.2. A u r o r a l E n v i r o n m e n t

257

integral number flux for energies greater than 14 keV is greaer than 108 cm 2 s ~ sr .....I. The highest potentials develop when there is a severe localized dropout of ion plasma density. This condition occurs more often during solar minimum conditions. One worst-case event was observed by the DMSP satellite on December 3 I, 1983 [19]. This event had the longest charging duration seen on any DMSP satellite, 62 sec. The spacecraft potential reached a peak value o f - 4 6 2 Vi. The measured parameters were as follows: Thermal ion density integrated flux of electrons Integrated flux or electrons (greater than 14 keV) Integrated flux (ion peak)

cm - 3 2.39 X 109 cm 2 sec 2.33 × 109 cm-2 sec- I sr 1.48 × lOs cm 2 sr 12.2

Fontheim [20] suggested that high-laitude precipitating electrons, which would be expected to influence spacecraft charging significantly, could be represented by the superposition of three distributions: a power law, a Maxwetlian, and a Gaussian, Analytic functions are easily manipulated to find the charging potential of a spacecraft and provide physical parameters that give. insight into the nature of the precipitating electron environment. The sum of three distributions is used to fit the energy spectra of precipitating electrons:

where ~p -

{Ap 0, E -~ ' I

-- E -EpL < < EpH E

.......4 / ~

(8.1)

4

~p represents a power law population linked m the energy of the precipitating p f i m ~ electron beam, composed of a Maxwellian distribution with temperature 0 describing the ambient flux for medium and high energies.. ~cj represents auroral enhancements that can best be described by a Gaussian distribution. Figure 8.3 shows a Fontheim distribution. Fontheim distributions for most environments have the parameters given in Table 8~6.

8. Space Environment and Interference

258 1010

10 9

_

_

--

L fit

10 8 -

107

......

.....................1 ............................

1 O0

t0

1 ................................................................... :.I: ............................................................. ,:) ,I ...............

10 3

1:04

10 s

: ..................1

i 06

Energy (eV)

F'~"e

8.3

Fontheim distribution for Ap = 3 × 1011 m '~3, a = 1.I, RpL = 50 eV, EpH = 1.6 × 106, n = 6 × 105 m -3, 0 = 8 keV, A o - 4 × 10 -3, E o = 24 keV, A = 16 keV.

T h e p a r a m e t e r s Ap and A o can be e x p r e s s e d t h r o u g h pp. and PG, w h i c h e x p r e s s the fraction: of cuffent containe~ in the p o w e r and G a u s s i a n p o t i o n

of the

distribution.: _

Jp

Vp Jp+Jm+JG'

=

Table 8.6 ~ p i c ~ Fontheim Distribution Parameters EEp EpH

A

Eo n kT ,.,. . . . . . . . . . . . . . .

Jo

PG J~+Jm+JG"

10-20 eV ! keV 2.54,5

i-I0 keV

5-15 keV 106--107 111-3 1-20 keV ::..:

..

,,,.,,,J..:

(8.2)

8.3. Effect of Electron Energy on Charging

8.3

259

Effect of Electron Energy on Charging

Figures 8.4 and 8.5 show how charging is affected by how energetic the environment is. Figure 8.4 shows the equilibrium potential for spheres in environments of various energies. For most materials, over most of the temperature range, the equilibrium potential increases roughly linearly with temperature. Some materials show a threshold effect, where little charging occurs until a threshold temperature is reached. The threshold effect seen here for some materials is an example of the threshold effect seen on all spacecraft,

Electron T e m p e r a t u r e (keY) 6

X

8

10

12

14

16

18

-10

IU

E

-20

:3

O"

m

-30

~0 ----P-----

Aluminum

D----Aquadg

Indox ...................... ; ..........=........:.=Screen

Kapton - - ~

Silver

.............., ..............C p a i n t

......".........M a g n e s i u m ---+---SiO

2

~

Gold

..............

Npaint

~

Solar

Teflon

Figure 8,4

F~uilibrium potential of a sphere in a geosynchronous substo~ as a function of substo~ energy for vinous materials.

8. Space Environment and Interference

2~

0.0

4

6

Hot Electron Temperature (keV) 8 t0

.

12

"-~-~

,,... + -~,~ +

v

ca

~C:

--tl.2

~ E

:~

'r~~. x

--..~. + ~

+ ...~

--tl,3 •

~

W

-4L5 ........................

...............................................................................................................

...... ". . .

Figure

:

:

:

:

:

:

:

:

:

Aluminum

Aquadg ~

Indox

Kapton

Screen ............ff:.:..~

:: .... : . •

-

~

Silver

:

:

:

:

:

.

~

==:::...==._~...:

, . _ ~ , _ . , _ . , . , , _ ~ . , . . . , _ , _ , o , ~ , _ . : , : . . : . , = = , : : = = . : = = . : . : . : : : : : , : . . :. . . . . . . . . . . . .

Cpaint ...........= ............... Magnesium

-.--+.---

Si02

~

Gold

..............e ..........................Npaint Solar

Teflon

8,5 Equilibrium.potentiai of a sphere in an aurora as a function of aurora energy for various materials. The environment is the severe auroral environment, except for the electron Maxwellian temperature, the upper and lower cutoff of the power law, and the energy about which the Gaussian is centered.

Figure 8.5 shows the equilibrium potential for 1-m radius spheres in auroral environments of various= energies. The equilibrium, potential increases slightly faster t h ~ linearly, and no threshold effect is seen in this range of energies.

8.4 Spacecraft Charging Effects Spacecraft surface charging is the buildup of net electric charge and therefore an electrostatic potential on the external surfaces of a space~raft due to incident

8.4. Spacecraft CharNng Effects

261

particles with energies in the kilo-electron volt to tens of kilo-electron volts range. A geosynchronous spacecraft charges when the vehicle encounters a region of enhanced plasma associated with a magnetospheric substorm. These enhanced plasma "clouds." have typical particle energies of 1 m 50 keX,: Large, lowaltitude, polar-orbiting spacecraft charge when they pass t~.~ough regions of auroral activity. Smaller spacecraft in low-altitude, polar orbits can charge because of multibody interactions, if they are near a larger spacecraft while passing through an aurora. Two types of spacecraft charging are of concern. Absolute charging is the development of a potential of the spacecraft frame relative m the surrounding space plasma. Differential charging is the change in the potential of one part of the spacecraft with respect to another. Differential charging may produce strong local electrical fields that can give rise to discharges. Spacecraft in geosynchronous orbit charge up to tens of kilovolts. The S C A T ~ satellite demonstrated that differential surface: charging on spacecraft during substorms is associated with discharges and. operational anomalies. In one event, potential differences of more than 9.5 kV were measured on the satellite [21]~ At the same time, 29 pulses were detected by the transient pulse monitor. Seventeen of the pulses exceeded the maximum instrument level of 7.4 V. Coincident with the discharges were three anomalies, including a 2-min loss of data. A survey of 9 years of SCATHA data shows a correlation between the current of panicles with energies in the tens of ~lovol.ts, the development of surface differential potentials in excess of 100 V, and electrostatic discharges [22]. A few severe charging events have been observed in the auroral region. During 1983, instruments on board the Defense Meteorological Satellite. 7 (DMSP 7) obse~ed an absolute potential of - 8 0 0 V [19]. Since then a few events with higher potentials, up to - 1 . 2 kV, have b u n observed.. No anomalies have been associated with any of the observed charging events. However, theory predicts that the larger spacecraft of the future will develop even higher potentials. Multibody interactions can cause or enhance surface charging if two electrically isolated spacecraft, such as the shuttle and an astronaut during extravehicular activity (EVA), are near each. other while in a hi.gh-energy (keV) plasma. Since multiple spacecraft have only be~n: flown in low equatorial orbits where highenergy particles do not occur naturally, charging due m multibody interactions has not been observed. As shown in Figure 8.6, surface charging causes problems for operational spacecraft. Differential charging can lead to significant potential differences between adjacent surfaces, and thus m discharges. The discharges are rapid pulses, typically of many amperes, for nanoseconds to microseconds.. A primary effect is the occurrence of electronic switching anomalies, which can be triggered

262

8. Space Environment and Interference insulating Room

Metallic Body _. --.

.......

"- N "" -~ "~ \ ~N~ "~ ,, \ " ~ " 7..~- \~ [ .~ ~,~ ~2 v

/

i / l~/" ~ "\

/ Glass . / / 4 = - - - Covered .,/" '~Solar

~Z' ~ v ~ ~;/~ ~ ZII~I ~

'~

~

N

/ \ . \

~

Protective Kapton Blanket IR Sensor

f

Particle Detector

~"--~\Can Antenna Have

EMI from Discharges

Figure 8.6 The effects of spacecraft surface charging include EMI, surface degradation, and contamination from discharges, disruption of particle measurements, and enhanced attraction of contamination.

by differential-charging-related discharges. The discharge-induced transients can cause system failures and, potentially, material damage. A more common anomaly is a phantom command, requiting inte~ention from the ground, possibly resulting in loss of data, thus shortening the operational lifetime of the spacecraft. Surface charging can cause increased levels of contamination, resulting in changes in surface characteristics. Spacecraft surface charging can enhance cont ~ n a t i o n in two ways. First, charged contaminants are attracted to oppositely charged surfaces. Some of the contarednants that would othe~ise drift away" from the spacecraft are attracted to the charged surfaces and impact at higher energies where chemical bonding is enhanced. Second, the material expelled during a discharge can be deposited on other surfaces. Contamination on surfaces with special properties, such as lenses, can destroy the special pro~rties. Higher temperatures may result from altered surface optical properties. Charging characteristics may change due to changes in secondary and photoelectron yields. Deposition of dielectric contaminants can also change surface conductivity. Finally, surface charging on spacecraft can bias plasma measurements of the space environment. The extent to which these effects interfere with the spacecraft mission varies from spacecraft to spacecraft and charging

8.4. Spacecraft Charging Effects

263

episode to charging episode. It was in the early 1970s that spacecraft began to experience anomalies, and in one case failure, that appeared to be spacecraftcharging related. The early 1970s is when computer-level logic in electronics subsystems was first introduced. The more sensitive electronics could be upset. by transients that did not affect the electronics on earlier spacecraft. As electronics become more sensitive, precautions become more. impo~ant. The process of charge accumulation on spacecraft surfaces is understood, and techniques have been developed to minimize the associated problems. NASA developed the Design Guidelinesfi~rAssessing and Controlling Spacecraft Charg~ ing Effects, which describes the understanding of the problem at that time and suggests techniques to avoid problems associated with spacecraft surface charg~ ing. Computer codes have been, developed to assist designers in the design of spacecraft with minimal surface charging effects. The first line of defense against differential charging is minimization of the area of surfaces that are insulators or floating conductors. This localizes the problem and reduces the amount of charge that can be rapidly discharged. (Sometimes the potential differences are larger when the areas are smaller, but the total charge and energy stored is smaller.) Careful attention m the design of the parts of the spacecraft where discharges are expected reduces the risk further. Shielding and filtering protect the circuitry from the EMI resulting from. any remaining rapid discharging. For some applications, the reduction of surface charging, both differential and absolute, either by using surface materials with high secondary electron emission (passive charge control) or by using a plasma emitter (active charge control) is necessary. Over the past 15 years, concern has arisen regarding charging on low-altitude, polar-orbiting spacecraft due m aurora precipitation. The 2-m DMSP spacecraft has been observed to charge to - i . 2 kV, and a 10~m spacecraft could charge m 10 kV. Auroral charging differs from. geosynchronous charging in that charging cu~ents tend to be much higher, the vehicle is in a charging environment for only s~onds, and the charging rate and the potential reached de~nd on the vehicle size. In addition, two-body and wake ef~cts can become important, and differential charging between vehicles such as a shuttle and an astronaut during EVA is of concern. The assessment of a low-altitude, polar-orbiting spacecraft design for possible charging-related problems requires the consideration of more complicated interactions and the use of different computational tools and environments than for geosynchronous spacecraft. In addition, low-altitude polar orbiting spacecraft need to work well while in the equatorial regions. The work of the early 1980s provided space~raft designers with tools to reduce the number and severity of surface-charging-associated anomalies, but left some questions unan-

264

8. Space Environment and Interference

swered. With the miniaturization of components, modem spacecraft are more vulnerable to EMI, so stricter requirements are needed.

8.5

T h e S p a c e c r a R as a ~ o a t i n g P r o b e

One way to understand the physics of spacecraft charging is to think of a spacecraft a Langmuir probe in its local., ionospheric plasma. The Langmuir probe (Ngure 8.7) is the most basic instrument used in laborat.o~ plasma experiments. It is used to measure the density and temperature of a plasma. Typically, it is a small metal sphere or long wire whose potential is swept through a limited range of voltages while the current to the probe is measured. The current is due to charged particles from the Nasma impinging upon the sphere. When the sphere potential is very positive compared to the kinetic energy of the plasma, only electrons are collected. When the sphere potential is very negative, only ions are collected. Between these two extremes there is a potential at which the ion current exactly balances the electron current, so that the current to the sphere is exactly zero. This potential, at which the net current is zero, is called the floating potential. Because, at a given energy, electrons move rapidly compared with ions, the floating potential is normally negative a few times the plasma kinetic energy. If the wire to a probe is cut:, the probe rapidly achieves the floating potential. In space, since there is no way for a continuous current to flow, the. plasma particles rapidly charge the spacecraft to a few times the electron energy. The difference between the laborato~ Langmuir probe and a spacecraft immersed in a magnetospheric s u b s t o ~ or an aurora is that the electron energies are a few volts in the laboratory and can be tens of thousands of volts in space. Laboratory floating potentials are typically negative a few volts in space; potentials ~ high as - 1 9 kV have been obse~ed. There are two models of current collection from a plasma. They are refe~ed to: as orbit limited and space-charge limited. Orbit-limited current collection is appropriate when the potential has a range larger than the largest impact parameter and is sufficiently well behaved so that no angular momentum b ~ i e r s exist. Potentials that vary more slowly than. with. the inverse of the radius squared satisfy these conditions. At geosynchronous orbit, the plasma is so dilute that little shielding occurs and the spacecraft potential drops roughly as the inverse of the radius. At lower altitudes where the plasma is denser, current collection is space-charge limited. The space charge of the attracted potential shields the attracting potential and thus limits the range of the potential. Ngure 8.8 illustrates orbit-limited current collection. In a high-energy, lowdensity plasma, the electron current exceeds the ion cu~ent and the vehicle

8.5. The Spececraft as a Floating Probe

265

! w";' pr°b;

Lit onl,y a, racts electrons. ..................

~i ............................... ::::::::::::::::::::::::::::::::::

@

................. =:--:

..........................

..................... :;;;;

.......

When probe is ve~ negative, it only attracts ions.

®@ Q

At some potential, the probe attracts an equal number of

electrons and ions,

If the wire is cut, the probe adjusts to the floating potential.

® ~

.

.

:

~

~.........

:_==__~

===__.=

:.-- ................. ==

......

®@ : -:-=-=-=:

=-=-=-.

::-=

.......................................

,,,.

.........

,,.,

....

:

..=-,..:

........

~.-.

............ , .................................................

=-:

....................... : . . . . . .

.~LLLLLLLLL'"'SSSSSSS:

Figure 8.7 A Langmuir probe attracts electrons and/or ions from. the su~ounding plasma depending on its potential.

charges negatively. As the potential becomes negative, the electron cu~ent diminishes because not all the electrons have the energy to overcome the potential.. If the plasma has roughly a Maxweltian distribution of energies, the electron current decreases exponentially with the negative, potential. Additional ions are attracted to the spacecraft as the potential becomes more negative. For the v e ~ low~density plasma in the magnetosphere, angular momentum limits the collection of ions. R e maximum impact parameter from which

~6

8. Space Environment and Interference

For a Maxwellian Plasma: no = density

0 = temperature

Vth

Repell~ Species Is Energy-Limited:

vl b

o ~ o 'for electrons Ie. = Ice Attracted S~cies Is Angular Momentum-Limited.: li = I ° ( 1 - ~ )

for ions

Potential for Current Balance:

bVth = avf

,(hi (vfl

I= o a

= io

Vth

Figure 8.8 Orbit-limited collection of ions on conversion of angular momentum.

ions are collected is that for which the ion's collected velocity must be tangent to the spacecraft to conse~e energy. The balancing of ion and electron, currents predicts a floating potential on the order of a few times the plasma temperature. Since the electron current diminishes exponentially and the ion cu~ent increases linearly, the principal effect of the potential is to decrease the electron cu~ent. Figure 8.9 illustrates space-chargeqimited current collection. Current col!ection by a spacecraft in a plasma with a Debye length, of the order of the spacecraft size is space-charge limited. As the spacecraft charges negatively, the additional ions collected shield, and thus limit the range of the potential.. To model, spacecraft charging in. an ionospheric plasma with. densities greater than about 109 m -3, space-charge-limited collection models should be used.

8.6

Charging

Environments

Two regions, geosynchronous altitude and the auroral regions, have plasma conditions where the plasma energy is high enough to charge spacecraft to kilovolts or higher. F i b r e 8.10 shows where these regions are located with respect to eart.h's size and the radiation belts. At geosynchronous altitudes, spacecraft charge

8.6. Charging Environments

267

For a Maxwellian Plasma: no = density

0 = temperature

Repelled S~cies Is Energy-Limited: I e = l ° e 05¢0 for

electrons

---/

ta

Io

Attracted Species Is SpaceoCharge Limited: 2

for,on Where. the Function f Has the Limits: f

a

()2 { rs

f~

1.08

~

keno a2]

I - Io a

2

- I°

(f)2

At the Noating Potential: ~e'/'/° = I° a Fibre

8.9

Space-charge-limited collection of ions based on conse~ation of angular momentum.

when enveloped in a "plasma cloud" injected during a magnet:ic substorm. These plasma clouds have p ~ i e l e densities of the order 1 0 6 t o 107 m .....3 and energies of I to 50 keV. For calculational purposes, measured fluxes can usually be fit by a Maxwellian or 2-Maxwellian distribution function. Under quiet plasma conditions, particles are of the order 10s m -3 with energies of the order 1 eV. Substo~s typically occur eve~ few hours, so the conditions for tens-of-kilovolts charging at geosyncl:~onous orbit occur frequently. The energetic electrons that charge low-altitude polar-orbiting spacecraft in the auroral region are those that generate the aurora boreaiis. While of similar origin to substorm electrons in the magnetosphere, the auroral electron fluxes can be as much as a hundred times as intense. Some of the enhanced intensity comes from the convergence of the magnetic field lines as they approach earth's poles. Measured fluxes can be fit well using the analytical form suggested [20]. Severe environments that can be used for design calculations are shown in Table 8.7. The environment to which spacecraft are exposed consists of more than the plasma environment. Neutral particles, electromagnetic radiation from the sun,

268

8. Space Environment and Interference

Auroral Region Magnetopause

©

\

Shock Front

Geosynchronous Altitudes

Figure 8.1.0 There are two regions where the electron energies can be kilovolts to tens of kilovolts.

high-energy charged particles, debris, and meteoroids all affect spacecraft. The atomic oxygen found at low altitudes can erode surface materials and affect the charging characteristics. Incident sunlight generates a photocurrent. Ultraviolet light can change surface characteristics. High-energy charged particles can deposit charge in insulators. This deep charging can interact with surface chaxging to generate discharges that would not occur if charge had not been deposited by both mechanisms [23]. Debris and met~roids erode surface coating. Atomic oxygen, debris, and meteoroids have densities of concern only at the lower altitudes.

8.7

Charging Currents

Spacecraft are designed for pu~oses other than acting as plasma probes. Consequently, the interpretation and prediction of the spacecraft potential are complicated because of the complex geometry and multiple surface materials, and the

8.7. Charging Currents Table 8.7

269

Severe Charging Environment

Geosynchronous Substo~ (Maxwellian for Each Species) Electron number density Electron temperature Ion number density Ion temperature Ion species

1.12 × 1.0¢~ 12 2,36 × I05 29.5 Hydrogen

m.-3 keV m-3 keV

Auroral (Cold Single Maxwellian for Both Species and Fontheim Electrons) Ion and cold electron number density Ion and cold electron temperature Ion species Energetic Maxwellian coefficient Energetic Maxwe!lian temperature Power-law coefficient Power-law exponent Power-law cutoff, low Power-law cutoff, high Gaussian coefficient Gaussian centered about Gaussian width

3.55 × 109 0.2 Oxygen 6 × t0 6 8 3 ×. !0 t~ t,I 50 1,6 ×. 106 4,0 x 1.04 24 16

m ~3 eV m -~s keV m -3 eV eV m -3 keV keV

absence of an easily accessible reference ground. Each insulating spacecraft surface interacts separately with the plasma and is capacifively and resistively coupled to the frame and other surfaces. Rather than a single floating potential, there can be a different one associated with each surface. Computing surface potential for a spacecraft is a considerably more complex problem than computing the potential on a conducting, spherical proM. As shown in Figure 8. I 1, currents other than incident electrons and ions should be included. Kilovolt electrons generate seconda~ electrons and can be backscattered (reflected) from surfaces. Kilovolt ions also generate secondary electrons. The current density of lowenergy electrons generated by solar UV emission is much greater than the natural charging currents. Because of this, most absolute spacecraft charging has been observed during eclipse, when the spacecraft is in the shadow of the earth. At equilibrium, each spacecraft surface is at a potential such that the net current to the surface is zero. The net cun'ent to each surface is INl~q" = IE --~SE-- IB -" II -- IsI - - l p -

1t,

(8.3)

2..70

8. Space E n v i r o n m e n t and Interference

• ~~

Electrons ............................ions

Photons Secondary,Backscattered, Photo Electrons Figure 8.11 Several currents contribute to the net current to a spacecraft surface.

where IE I~ IsE IsI IB ./p 11

= = = = = = =

electron current to surface ion current to surface secondary electron current due to I E seconda~3, electron current due to I ! backscattered electron current due to I E photoelectron current current to adjacent or underlying surfaces.

Ea:ch of these currents is a function of the spacecraft geometry and velocity and the plasma conditions. To get a feel for each of the t e ~ s in this equation, consider the current to a negatively charged isolated sphere in a Maxwellian plasma in the orbit-limited current collection regime. The electron current is given by

/

I~ - en ~/ ~

e0

e

~o

,

(8.4.)

where e is the electron charge, n is the plasma density, 0is the plasma temperature, me is the electron mass, and ~ is the surface potential. The ion current is given by

li _ e n ~//~eO where m i is the ion mass.

\

- ~l'

(8.5)

8.7. Charging Currents

271

The interactions of the incident etectmn and ions with the spacecraft surfaces have a. profound effect on floating potentials. The most important process is secondary electron emission [241. Because secondary electron yields are so high for many surface materials, the spacecraft floating potential is often positive! As se~n in Figure 8.12 for electrons with energies between 50 eV and 2 keV, more than one secondary electron, is emitted for every incident electron from a material such as kapton. This results in a positive charging current. Only when the electron energies exceed several thousand volts does the spacecraft charge negatively. Backscatter yields are less than unity and vary little with energy. Ion-generated secondary electrons enhance the ion= current and. act to reduce absolute charging levels. The photoelectron current for sunlit surfaces is of tlne order of 2 - 4 × 10"sA m 2 for most spacecraft surfaces. The current m the underlying and adjacent surfaces depends on the surface and bulk conductivity and the geometry for additional. information on these calculations. As the net current to the kapton sphere is zero at - 2 2 kV, the floating potential is - 2 2 kV. As is shown in Figure 8. t 3, at low applied potential, the electron current, which drives the charging, is 3 × I0 .....6 A m ......2, half of which is immediately canceled by secondary and backscattered electrons. As silver generates more s~ondary and bac.kscattered electrons, the

E. Secondary

.~ >.

~

I. Secondary

Backsca~er

2

0

0.01

Figure 8.12

0.1

1 Energy (keV)

t0

1~

Electron-generated seconda~ electron yield, backscatter yield, and protongenerated secondary yield for kapton.

272

8, Space. Environment and Interference

...........................................J T O T

JE

-----Z3r----= J S E C E

<~

..........~.... J B S C A T

~-2L)

JI

-3

..............*

.•

.

-30

.

.

.

.

.

.

I

|

-20

I

"

,

| ....................................

-10

--

JSECI

i

0

Potential ( k V )

Figure 8,13 Current vs voltage for a kapton sphere in a severe substorm environment. The floating potential is -22 kV.

silver sphere's equilibrium, potential is lower. Ion-generated secondary electrons effectively triple the incident ion current. Because of the secondary and backscattered electrons, current balance is effected equally by diminishing the electron and increasing the ion currents. Because the incident electron spectrum remains Maxwellian, electron-generated secondaries and backscattered electrons, remain a constant fraction of the incident current as the spacecraft charges. R e iongenerated secondary electrons increase compared with the incident ion current because the energy of the ions increases as the spacecraft potential, becomes more negative. Ion-generated secondary electrons yield peak ion energies of several tens of kilovolts. The different conditions of substorms and aurora means that the important contributions, to the net current in these two regions are different. Figures 8.13 and 8.14 show that charging of geosynchronous spacecraft is dominated by the balance of the incident electron current with the secondaxy e!ectmns (from incident electrons and ions). Figure 8.15 shows the current vs voltage for the various currents that contribute to the charging of a l-meter kapton sphere moving at

8.7. Charging Currents

---------

?

JTOT JE

o

E <

..........................J S E C E

C~

0

273

JBSCAT -2

-

• JI

JSECI

-3

- 4

, - | ~ = = : = = = -

~

====-==~

-30

|

........

-20

-10

0

Potential (kV)

Figure 8,14

Current vs voltage for a silver sphere in a severe s u b s t o ~ environment. The floating potential i s - 5 . 8 kV.

0 -....................... =-~ .

...............

?

E

~ ~

.

.

.

.

:

: . . : .

LLLLLLLLLLLL :.:~

....

. IT(3T

...........................

..........JE

20-

---~'--~JSECE

............ JBSCAT

o

O~ j S E C I ill

-20 ...

..................................................... i

.

.

.

.

.

................

1~. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

-0.8

1_ . . . . . . . . . . .

|

) ...............................

11

-0.6

-0.4

, ,i ..................................

I . . . . . . . . . . . . . .

-0.2

............... ==--=':

...............

)

!

0

Potential (kV) Figure 8,15

Cun'ent versus voltage for a 1.-m kapton sphere moving at Mach 8 in a severe auroral environment. The floating potential i s - 2 3 0 Vi

274

8. Space Environment and Interference

Aurorat/EI/e/ctrons

Substorm Electrons

•

..

,!:!> ~

-S' econdary Electrons

Figure 8,16

~S~condary Electrons

...................~.... . . . . . . . . ,-

The charging of GEO spacecraft is determined by the balance of the substo~ electrons and the secondary electrons. The charging of large spacecraft in the auroral region is dete~ined by the balance of the net auroral flux and the space-charge-limited ion flux.

Mach 8 (orbital velocity in low-eaCh orbit) in the severe aurora environment. The dominant currents are the space-charge-limited ram ions and the incident electrons. Figure 8.16 illustrates the difference between these two regimes. Char& ing of large objects in low-altitude, polar orbit is determined by the balance of the net aurora flux and the space-charge-limited ion flux. These spacecraft leave a substantial ions-depleted wake, The lower ion density in the wake region means fewer ions are available to neutralize the built-up negative charge. The comparison (shown in Table 8.8) between the potentials calculated ignoring space charge with those calculated including it show more than an order of magnitude difference. The space-charge-limited result agrees with observation.

Table 8.8 =-:

=-:.

,_,

:u::

.....

-=-=-=-=...

Equilibrium Potentials Calculated by Space Charge ,, =,,,,,,,,,,,,,,,,,,,

:=:

:

............... ,,.

....................

:::::::::::::::::::::::::

=:::::::

:

...............

Orbit.limited Sphere radius Mach velocity

::,_....

......................

-9 V -5 V ...............

:;:;:::: ...............

: :

.................

1m

Kapton Silver ..,,

: ................

....................................

---=-=-=-:

.: .:.

:. :. :......,=c.:::::::::::::::::::::::::::::::::::::::::::

==

Space.Charge-Limited

::,,,.:

i0 m

0.~1

8

0.~1

8

-550 V -250 V

-230 V -99 V

-5400 V -3100 V

-29~ V -16~ V

.....................................................

.

.

.

.

.

.

.

.

.

.

.

.

.

.

.....

-

..........

.:::::

_.,o

.............

.:::::

...................................... -

-

-

-

8.8. Differential Charging

8.8

275

Differential Chargng i

The electrons associated with surface charging penetrate tess than a micron into the spacecraft skin. Because of this, surface coatings play a large role in determining spac~raft potentials. While the time to achieve net current balance is very short, on the. order of a millisecond, the time for each surface to achieve its own equilibrium potential is thousands of times longer. The. development of differences between the potentials of different surfaces is referred to as differential charging. Figure 8.17 shows mechanisms for the development of differential potentials. Different materials have different equilibrium surface potentials because the secondary and backscatter yield coefficients are different. This effect can lead to potential differences between neighboring surfaces. The ground, potential of the spacecraft depends on the average prope~ies of the spacecraft surfaces. Each surface charges differently from spacecraft ground. The difference between the surface, potential of a dielectric and the potential of the underlying conductor is another source of differential pomntials.

Figure 8.17 Differential potentials due to difference in secondary emission properties.

276

8. Space Environment and Interl~rence

The rate of absolute charging is determined by the capacitance of the spacecraft to infinity, while the rate of differential charging is determined by the capacitance of the dielectric layer. The rate of change of the potential on a sphere is given by d_~ = dt

!

1

4 ~o R

=

JR

= e0 '

(8.6)

where I is the current to the sphere, J is the current density to the sphere surface, and R is the sphere radius. For a 1-m sphere and a net current of 1 # A m -2, the sphere begins to charge at a rate of 1~ kV sec -~. As the potential on the sphere increases, the net current decreases, as can be seen in Figure 8.I3. Therefore, the charging rate decreases with time. The rate of change of the difference in potential across a dielectric layer is given by d~.: = J d dt

(8..7)

eo '

where d is the thickness of the layer. A 100~#m layer (4 rail) with the same incident current density charges at a rate of ! 0 V sec .....5 Therefore, differential. charging takes place 1.04 times more slowly than absolute charging. Figure 8.18 shows the equilibrium potential contours calculated by the com~ puter code NASCAP of the N A S C A P / G E O codes for a silver quasi-sphere of radius 3.5 m with a 5-rail-thick kapton coating over half of the surfaces exposed to the severe substorm environment of Table 8.3. The surfaces with the kapton coating are those facing the bottom, the left, and the rear. Figure 8,19 shows the time history of charging of this sphere, In 1 sec the entire sphere charges to - 18 kV. After 1500 sec (25 rain), a differential potential of I kV has developed. By 1~,000 sec (28 h), the equilibrium potentials of -22.51 kV on the kapton and - 1 0 . 9 kV on the silver are reached. A thinner kapton coating would charge more. slowly, Photoelectron current densities are about. 2 - 4 ×. 10s A m °, typically an order of magnitude greater than incident electron currents, even before secondaries are taken into account. As a result, in sunlight, high absolute potentials are rarely observed on spacecraft. However, in an intense substorm, spacecraft shadowed from the sun slowly charge to thousands of volts negative, while the sunlit surfaces remain a few vol~ positive. This. process can lead to differential charging as illustrated in Figure 8.20. High. differential potentials can develop between shaded and unshaded dielectric, and high differential potentials can develop between shaded dielectric and spacecraft ground. The sun-shade interface, and

8.8. Differential Charging

277

t7 16 15 14 13 12

11 •~ x <¢

10

2

3

4

5

6

7

8

9

10

11

12

13

14

15= 16 17

X-Axis

F i b r e 8.18

A 3.5~m radius quasi-sphere, half kapton and half silver, exposed to a severe geosynchronous environment develops different potentials on the surfaces coated with different materials. The contour levels are at l-kV increments. The kapton surface charges to -22..5 kV, and the silver surface charges to 10.9 kV.

8. Space. Environment and Interference

278

T i m e (see)

10--3 0~

0,1

10

10 3

10 5

1:07

~< -10 :>

v

m

m

C

d~ -15

~0 -2:5

I

III •.

_'..

Kapton

~

Silver

Figure 8,1.9 Time histo~ of the potential of quasi-sphere shown in Figure 8.18.

Sunlight

S S S "-~',,,...

Photoemission

+

+

+

+

+

-

L

phot~mission in shaded regions

No

Figure 8,20

Differential charging between shaded and unshaded surface in sunlight.

8.8. Differential Charging

279

the differential potentials, shifts location through the year. A sun angle that does not cause a problem in one season can in. another. Sunlight differential charging can lead to. high spacecraft ground potentials when the low-energy photoelectrons cannot escape from the spacecraft because of potential barriers [25, 26]. Typically, the saddle point is driven by the balance of photo and secondary electrons and has a height, of a few volts. All the surface then charges negative at a rate co~esponding with differential charging, typically a few hundred volts per minute. Figure 8.21 shows the time histou of the ground potential and the surface potential on the sunlit and d a n sides. The sunlit side st~ts out a few volts positive. The dark side charges negatively. After a half second, the dark side is at - 8 V and the sunlit side at 3 V. A potential barrier of 1 V has developed. The potential bamer grows until the photoelectrons cannot escape. Once the photoelectrons cannot escape, the entire space charges. Sunlight charging is a multidimensional effect calculable only by models that include the three-dimensional geometry of real spacecraft. As illustrated in Figure 8.22, during aurora, low-altitude, polar-orbiting spacecraft can develop differential potentials between their ram and wake sides. The ram side of the spacecraft has a much higher ion current than the wake side,

Time (sec) 0.01 ,

1

t00

104

t06

-5 -t0 -15 -20 -25

i

I

~

Sunlit

"-

Ground

I I

Fig~Jre 8.21 Time history of surface potentials and spacecraft ground on the sunlit kapton quasi-sphere.. Charging of spacecraft ground occurs on the differential charging time scales due to the formation of a barrier: The final ground potential of -18 kV is reached after I ~ 0 sec (17 rain).

280

8. Space Environment and Interference

Ram Ions ............_=_=_=_=~ _=_== .........................................= _ = . _ = _

............/~..

................................................................ ~._

Beam

J=~ 11~

Auroral Beam

Figure 8.22 During aurora on low-altitude, polar-orbiting spacecraft, differential potential can develop ~tween the ram and wake sides of a spacecraft.

which must pull in the ions around the spacecraft. The ions reaching the wake side of the spacecraft are attracted from the sheath edge. In this calculation a low-potential region develops on the wake side of the sphere because of the focusing of ions due to the high symmetry. The ground potential is - 6 5 0 "~ the surfaces in the ram are at - 1 V, the surfaces in the center of the wake side are at - 1 : ~ V, and the peak potential on the w ~ e side is - 8 4 0 "v'~The same quasisphere with a Mach velocity of 0 has an equilibrium potential of - 1 . 9 kV.

8.9 Arcing The primary mechanism by which charging disturbs spacecraft is through discharges. The rapid discharging of surface, or arcing, can disrupt operations, disturb measurements, and damage instruments or spacecraft surfaces. The mechanism for triggering discharges is an active field of investigation. Measurements made in the laboratory produce results that cannot be extrapolated to on-orbit spacecraft. Part of the problem is the impossibility of adequately simulating the space environment, the spacecraft, and their resultant interaction. Some e x ~ p l e s of this inadequacy are the breakdown thresholds, area scaling, and satellite response. It is observed that in space, discharges occur on spacecraft when the calculated and measured potentials and electric fields are lower flaan those needed

8.9. Arcing

281

to generate discharges in. typical electron-beam laboratory experiments. Measure~ ments made on uniform, small dielectric samples find that the discharge current scales directly with the square root of the sample area, and the fraction of charge blown off by the discharge is essentially constant for each dielectric type [27]. Large sample measurements that include lapped and butted seams of materials present in real spacecraft, thermal-control surfaces differ from the uniform, small-. sample results [28]. Even though the present understanding of discharges is not. complete, there are some well-established data, plausible analytical treatments, and useful criteria that can be employed in. predicting the location and frequency of discharge. Flashover discharges occur when a layer of neutral gas atoms that has been generated by electron-stimulated desorption breaks down under the local electricfield, stresses caused, by differential, potentials of spacecraft dielectric [29]. This type of discharge frequently occurs at the edge of a dielectric next to another surface, at cracks in dielectric exposing conductors underneath, or at exposed conductor-cover cell interfaces in solar a~ays. The typical field strengths required for flashover is 2 × 10('V m ......t [29], which is an order of magnitude below that for dielectric breakdown, Breakdown of dielectric also provides discharge paths for spacecraft, dielectric. Dielectric breakdown results from the bulk failure of the dielectric material. Two of the theories of dielectric breakdown are (1) that the energy stress produced by the charged surface potentials exceeds the binding energy of the molecules that make up the material, causing rupture of the bonds and loss of the material; and (2) that field-enhanced electron emission provides electrons that generate cascades and heating until material vaporizes. Dielectric failure most often occurs at imperfections or dielectric weak points.. Dielectric failure at an imperfection usually appears as a pinhole through the dielectric and is referred to as punchthrough. Localized material breakdowns are also observed in trapped charge. layers of dielectric about 1 ~m below the dielectric surface. These bulk material failures appear as channels or tunnels just below the dielectric surface and are accompanied by surface damage where the channels penetrate the surface. The typical field strength of dielectric is 2 × 107V m I. B lowoff is a large-scale discharge phenomenon occurring at the surface of the dielectric and sometimes extending over the entire vacuum-dielectric interface. Several. models have been proposed for the primary mechanism that initiates blowoff discharges. One of the most promising is a surface discharge model [30]. The essential idea of the Stettner-DeWald model is that the intense: electric field on the boundary of the charged and discharged region of dielectric surfaces accelerates ions into the surface. Ions with the necessary velocity and angle to

282

8. Space Environment and Interi~rence

the surface cause kinetic emission of electrons, resulting in discharge of the surface. The incident ions also cause sputtering of surface atoms and ions. Newly created ions are also accelerated into the surface, producing further discharge and movement of the charge/discharge b o u n d ~ ~rther into the dielectric's charged region. The initial ions for the Stettner-DeWald model may come from the ions in the ambient plasma or ions desorbed by the incident electrons. A variation of the Stettner-DeWald discharge model, termed the localized plasma sheath model, has been proposed by Kxauss [31] to overcome some of the shortcomings of the ion surface-discharge model. The ion surface-discharge model and variants are promising in that they explain most of the observed features of laxge-scale dielectric discharges and are supported by surface-discharge measurements. Unfortunately, a determination of their validity and usefulness for determining where and how often blowoff discharges occur require further theoretical and experimental work.

8.10

Determination of Path for Discharge Energy

The ~ansient pulse produced by the discharge couples directly or capacifively to the spacecraft structure and to spacecraft cables. Various methods are available to analyze the coupling of the discharge transient to the spacecraft elements. Some of these different coupling analysis methods are briefly discussed next. &lO.1

LUMPED-ELEMENT METHOD

Lumped-element modeling (LEM) is the most cormnon analysis method used for discharge coupling. LEM models replace spacecraft structur~ elements with their equivalent inductance, capacitance, and resistance. They have v ~ e d from simple one-dimensional models, where a few circuit elements are used, to models of the entire spacecraft structure [32], to very complex three-dimensional models where each structural element is modeled by its equivalent RLC circuit [33]. The discharged source is then used to drive the lumpeA-element electrical model at the point in the model associated with the cells identified as locations where discharges are likely to occur.. A circuit analysis code such as SPICE is used to analyze the lumped-element circuit model and to calculate the current flowing on each structural element due to the discharge source. A LEM model is also generated for cables that are attached to or run new the spacecraft structure. The structure cu.~ents calculated by SPICE for the spacecraft LEM are then used to drive the cable LEMs. SPICE runs are made for the cable electrical models using

8.10. Determination of Path for Discharge Energy

283

the structure drive sources to determine coupling of the structure, cu~ents to the cable conductors. The conductor cmTents and the resulting voltage appearing at interface circuits attached to the cable conductor are compared to the interface circuit thresholds to determine if the interfaces will be upset or damaged by the discharge sources. &lO.2

NUMERICAL ELECTROMAGNETIC METHOD

The numerical electromagnetic method uses codes such as the Numerical Electro° magnetic Code [34] m analyze discharge coupling. For NEC the spacecraft structures and cables are modeled as a combination of "rods" and flat polygonal "plates." The NEC spacecraft model is then. driven by direct sources representing the discharge at the model locations identified with the cells that are likely to discharge. NEC is a frequency-domain code, and thus the time-domain discharge sources must be c o n v e g ~ to the frequency domain when. input to NEC. The NEC program solves both an electric field integral equation (EF1E) and a magnetic field integral equation (MFIE) and accounts for mutual coupling between spacecroft model elements. Individual NEC calculations are performed at a single frequency point and thus must be repeated at a number of specific frequencies that cover the range of interest for the frequency portion of the discharge sources. The NEC code output is transfo~ed to the time domain to yield the current flowing on. the cables of interest. The cable currents are then used to determine the cable conductor currents via the transfer characteristics of the cffNes being analyzed or with a iumped~element electrical model. Once the conductors' currents are found, the voltages and. currents appearing at sensitive interface components are dete~ined using usual network analysis techniques. 8,10.3

PARTICLE-PUSHING METHODS

~ e . particle-pushing methods use system-generated electromagnetic pulse (SGEMP) codes such as the two-dimensional Arbitrary Body of Revolution Code (ABORC) [35] or its three~dimensional equivalent MEEC to calculate spacecraft discharge response. These codes solve Maxwell equations by direct finite differencing for asymmetric geometries. Spatial cun'ent densities are obtained from finite particles of charge that are followed through the spatial mesh of zones. In addition to the Maxwell. equation routine, ABRC has a Poisson equation solver from which electrostatic field may be obtained for each time step. Woods and Wennas [36] describe the details of using ABORC to perfo!~ discharge, analysis of spacecraft. Discharge response calculations begin with the static: fields arising

284

8. Space Environment and interference

from the initial charge on the dielectric. The ABORC discharge model assumes uniform spatial emission, a triangular time history, and zero initial electron kinetic energy. Flashover currents are included to the extent that the potential across the dielectric varies with time due to both blowoff and flashover effects. Reasonable agreement has been found between discharge results calculated with ABORC and actual measurements made on spacecraft models in the laboratory.

8.10.4

EMC ~ D I A . T I V E COUPLING METHOD

The EMC radiative coupling method employs EMC analysis codes such as Intrasystem Electromagnetic Compatibility Analysis Program (~MCAP) and Specification and Electromagnetic Compatibility Program (SEMCAP) to analyze discharge coupling for spacecraft. IEMCAP and SEMCAP contain communications, and EMC analysis math model.s to efficiently evaluate the spectra, and transfer modes of electromagnetic energy between generators and receptors within a system, in analyzing a system with these codes, all. system emitters are. characterized by emission spectra and all receptors are characterized by susceptibility spectra. All ports and coupling mechanisms are assumed to have linear characteristics. E~ssions fi'om the various emitter ports are assumed to be statistically independent so that signals from several emitters, impinging at a receptor poR, combine on an. RMS or ~ w e r basis. The function of these codes is to dete~ine, by analysis, whether the signals from one or more emitters entering a receptor port cause interference with that receptor: Electromagnetic interference (EMI) is accessed by computation of an EMI margin for each receptor port. The EMI margin is just the ratio of power received at each receptor port to that receptor's susceptibility. Coupling models built into these codes include antenna coupling, wire-to-wire coupling, case-touche coupling, coupling through filters, and fieldto-wire coupling.

&lO.5

RECOMMENDED COUPLING ANALYSIS APPROACH

The present standard practice used for spacecraft discharge response is the LEM method. The analysis procedure is as follows: i. Obtain the stractural details of the spacecraft m analyze. 2. Obtain the physical details of spacecraft cabling that is routed on or near stmcturai elements and connects to sensitive interface circuits that are susceptible to interface or damage.

8.10. Determination of Path for Discharge Energy

285

3. Construct a physical model of the spacecraft using the following library of structural components: rectangular plates, disk plates, hollow cylinders, bars, and cones. 4. C o n s ~ c t an electrical model of the spacecraft replacing the structural components with their equivalent electrical RLC models. The equiva, tent circuits for plates, hollow cylinders, and disks are shown in Figure 8.23. The structural elements fall into one of two major categories, the exterior structural elements that are capacitively coupled to space and the interior elements that capacitively couple to each other. These free space, capacitances and mutual capacitances are included in the electrical model for each model structure. Capacitances and inductances of stracture

Lser Rser ~

[

I

J

'

~

HOLLOW CYLINDER

PLATE. ......... - "

................ ~..................................... ~ .................... -

Lser Rser

DISC

Figure 8.23 Equivalent circuits for plate, cylinder, and disk.

286

8. Space Environment and Interference

elements may be found in Granger and Ferrante [37]. The eleztrical model nodes should co~espond to structural features with dimensions of 0.3 m or less. 5. Construct LEM electrical coupling models of the spacecraft cabling and insert these cable coupling models at the appropriate locations in the spacecraft electrical model. Simple LEM models can only be used for cables that are less than M10 long, where A is the smallest wavelength used in the analysis. For longer cables, distributed LEMs must be used, where the simple LEM is divided into several sections to account for distributed effects, 6. Insert the discharge sources at the appropriate points in the spacecraft electrical model. The sources drive the spacecraft electrical model through their source impedance, which is capacitive or resistive de-. pending on their spacecraft injection mechanism, Use a circuit analysis code such as SPICE to solve the spacecraft electrical, model for the voltages and currents that appear at the cable conductor loads representing input impedances of the sensitive interface circuits.

8.11 Circuit Upset Circuit upset is a nonpermanent alteration of a circuit or component operational state that is self-correcting or reversible by automatic or manual means. Some examples of upset are provided in Figure 8.24. The conditions under which upset occurs when a circuit is, stressed by a discharge transient are as follows: 1. The dischm'ge transient's amplitude must be a significant fraction of or greater than the circuit's operating signal levels. 2. The discharge tra.n :~ent s time scale must be within the circuit's response time. •

S"

~

.

3. When the discharge transient's time scale is shorter than the clr rats response time, the discharge transient's amplitude required to cause upset exceeds the circuit's operating signal levels by increasing amounts as the time scale differences become larger. ~

C

~

'

4. For digital logic circuits, when stressing discharge transients have time wi.dths that are within, the logic circuit's response time, logic upsets occur when the discharge transient's amplitude is greater than the logic c!r mts noise margins.

8.11. Circuit Upset Discharge

I d Input

I~1':~A'ransient ~

. . . . . . . . . . . . . . .

j

:

K

...................

FlipFlop

Q

•. ........

~

|

Q

287

/ o-output | 'Low

11!

__.

_Jl=.

1

~ - Flip-Flop Response Delay Time "

ii

.: II .....! ..........................I..i

..................~ ,

. . . . . . . . . . ._~ .......

"High"

(a) Flip-Flop Upset Power S .

upply Input

Power Supply

......

.................... ~ ................

Discharge ~.Transient

i b, [ .... ~./~. ~¢ v l ,iv r~ _ . ....................... r ........... ,._ 1

::

I I

4t-] i-~ Gate Response Delay Time C-Output i I"High" ..................................... II

(b) NAND Gale Upset f Adnput

c

B

C-Output

f

Discharge

Transient

Amplifier Response Delay Time

[ ~ i l I1 I..~; ~ • • " '~1 I Ill 4 1 IIi ii I t, :I" ......~ " : J

",J \J

/-'~q

2 ;2 == 1

U V \~

•

(c) Amplifier. Upset

Figure 8.N

Examples of upset from discharge transient pulses.

The upset thresholds for representative logic families are given, in Table 8.9. The upset levels (e.g., noise margin) for commonly used logic families vary from a few hundred milli-electron volts to a few volts. Typical upset energy level thresholds range from l to 50 nJ.

288

Logic Family

Power Supply (V)

Typical Gate Quiescent Power Dissipation (mw)

Typical Upset Threshold and Characteristics of Some Logic Families Typical Propagation Dekzy (nsec)

Typical Signal Line DC Noise Immunity (V)

HTL SCL CMOS CMOS CMOS

5 5 15 5.2 5 10 15

5 15 30 25 0.000025 0.00010 0.00023

30 10 85 2 45 16 12

Logic Voltage Swing

(v)

Typical Energy Noise Immunity on Signal Line (joules x I @ )

High

Low

DTL TTL

Z)pical Signal Line Impedance (ohms)

Min

Z)p

Min

Typ

Low

High

0.7 0.4 5.0 NA 1.5 3.0 4.4

1.2 1.2 7.5 0.2 2.2 4.2 6.5

0.7 0.4 4.0 NA 1.5 3.0 4.5

3.8 2.2 7.0 0.17 3.4 6.0 9.0

50 30 140 7 600 300 250

1.7K 140 16K 7 12K 600 450

Low 4.5 3.5 1.3 0.8 5.0 10.0 15

3

High

3

15 25

48 NA 3 10 22

NA 15 5 13

-

8. Space Environment and Interference

Table 8.9

8.12. ComponentDamage

289

8.12 Component Damage Component damage is a permanent change in one or more electrical characteristics of a circuit component. Circuit components are vulnerable to thermal damage and electrical breakdown when stressed by dielectric discharge transients. The damage energy threshold for various circuit components for a I ~ - n s e c rectangu1~ transient pulse is shown in Figure 8.25. R e damage threshold level ranges from I0 nJ for microwave diodes to several hundred nanojoules for various logic families.=

Component Transformers

Switching

Diodes

Zeners

owo,

Microwave Diodes LED

Transistor

Signal •

.

r

Linear IC ITL s/c

._rtr

jFET

MOS CMOS/SOS Capacitor

Voltage Dependent Failure

~ ~

Thin Film Resistor

Open o---~v,---o [ 20*/o Resistance ° - - A v " ° Charge

Carbon Resistor Relay

o - - J ' --o Weld Contact 10-10

10~

I

10-2

1

102

. . . . . . . . . . . . .

106

Energy (J)

Figure 8.25 Permanent damage energy threshold of components for lf~-nsec pulse.

290

8. Space Environment and Interference

For semiconductors, the most common discharge transient damage mechanism is l~alized thermal runaway triggered by electrothermal overstresses. ~ i s condition produces a resolidified melt channel across the junction once the transient is removed, where the melt channel appears electrically as a low-resistance shunt across ~ e junction. Junction damage is most likely to occur when the discharge transient reverse biases the junction and drives it into second breakdown. Forwardstressed junctions also fail, but typically have damage tbxe.sholds that are 3 to 10 times higher than reverse stressed junctions. For integated circuits, metallization burnout and gate oxide breakdown (for MOS devices) are also prominent failure mechanisms. Semiconductor failure thresholds for discharge transients can be predicted from ~ o w n or measured data using models developed for discrete semiconductors and integrated circuits. Tinese models, which are based on thermal considerations and experimental, results, yield an expression for the failure threshold level, (8.8)

Pp = k~t ~/2

where Pp is the. power in. watts required in t seconds to produce device failure, and k t and k2 are device-dependent damage constants. As illustrated by Figure 826 for discrete devices, k2 is unity for discharge pulse widths less than 100

PF

tp -1

t.p-1/2

tp0 |

~

i t i I t

ta

te

Log tp

Figure 8.26 Pulse power failure dependence on pulse width for discrete semiconductors.

8,12. Component• Damage

291

nsec, 0.5 for pulse widths between 100 nsec and 3 ~ msec, and zero for longer pulse widths. The value of k~ is determined by test when. possible. Measured values of k~ for some common discrete semiconductors are shown in Table 8.10. When test data is unavailable, the value of k~ can be obtained from data sheet information and the analytical expressions given. The failure models for diodes and transistors are shown in Figure 8.27. The diode and transistor junctions am modeled by a resistor that represents the junction's bulk resistance and a voltage source, that represents the reverse bre~down voltage for the junction. Typical. values of junction bulk. resistance and reverse breakdown voltage for diodes and transistors are listed in Table. 8.11. For integrated circuits, k~ and k2 are determined experimentally when possible. For the c ~ e where test data is not available, typical values of these coefficients for diffe~:ent types of inmgrated circuits have been determined by tests and are given in Table 8,12. Tight integrated circuit manufacturing tolerances and standard circuit, designs have allowed integrated circuits to be grouped by their technology into generic failure classes and their terminals categorized into one of the following types: input terminal, output terminal, and power mrminal. The IC terminal failure model consists of a resistor representing the terminal's bulk resistance and voltage source representing the t e ~ i n a l ' s reverse breakdown voltage.

Table 8.10

Damage Constants and Junction Breakdown Voltages for Some Typical Discrete Semiconductors

Device

Type

1N750A 1N756 1N914 1N3600 IN4148 IN4~3 2N918 2N2222 2N2857 2N2907A 2N3019 2N3440

~ner Zener Diode Diode Diode Diode Transistor Transistor Transistor Transistor Transistor Transistor

"kl = k, k 2 = 1./2~

~

( W s t/z)

2.84 20.4 0.096 0,18 0,011 2.2 0.0086 0,11 0,0085 0. I 0A4 1.1

B V ~ ° (V)

B Vcn o (V)

V,D (V)

4,7 8.2 75 75 75 200 3 5 2.5 5 7 7

30 60 30 60 140 3~

292

8. Space E n v i r o ~ e n t and Interference

Transistor Damage Model

PF = kt, t~k2

+

IF =

C

4RBPF 2RB

V F = RRIF + VBD

IC Terminal Damage Model Power

E

input

Output

C :.::::

Diode Damage Model

=-:0

VF--~ li

Anode

RB VBD

Cathode

Reference

Figure 8.27 Transient pulse failure models for ~ansistors, diodes, and integrated circuits.

Table 8.11 Typic~ Junction Bulk Resis~nce for Discrete Semiconductors .

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Device Category

Reverse Bias (ohms)

Forward Bias Whms)

Zener diodes Signal diodes Rectifier diodes Low-frequency transistor (e-b) High-power transistor (e-b)

1.0 25.0 150~0 10.0 2.0

0. I 0.25 0.05 1.0 02

8.12. Component Damage Table 8.12

Damage Constants and Failure Parameters for Va~o~ Logic Families

Category

Farefily Terminal TTL RTL

DTL

ECL

MOS

Linear

293

Input Output Input Output Power Input Output Power Input Output Power Input Output Power Input Output

l~wer

Upper

K1

Kz

"v'Bo (V)

R B (ohms)

95% K t

95% K z

0.00216 0.00359 0.554 0.0594 0.0875 0.0137 0.~40 0.0393 0.I52 0.0348 0A56 0.0546 0.001.4 0.105 0.0743 0.0139

0.689 0.722 0.384 0508 0.555 0.580 0.70.6 0.576 0.441 0.558 0.493 0.483 0.819 0.543 0.509 0.714

7 1.3 6 5 5 7 1 1 20 0.7 0.7 30 0.6 3 7 7

16 2A 40 18.9 20.8 25.2 15.8 30.6 I5.7 7.8 8.9 9.2 i 1.6 10.4 13.2 5.5

0.00052 0.00098 0.12 0.0096 0.026 0.0046 0.012 0.009 0.~5 0.0031 0.22 0.0063 0.~2 0.038 0.0054 0.0045

0.00896 0.013 2.6 0.39 0.70 0.041 0.0136 0.17 0.51 0.397 0.935 0.47 0.0046 0.29 1.01 0.043

From the failure models for discrete semiconductors and integrated circuits, we find that the failure voltage VF and failure current I F at the device t e ~ n a l s are expressed as V ~ = V B + iFR B

I F, = (VB - ~ , ~ _

(8.9) 4RBPp)/2RB,

where VB is the t e ~ n a l ' s breakdown voltage, R B is the terminaI's bulk resistance, and Pp is the terminal's failure power, given previously. The conditions for failure to occur when. a circuit is stressed by a discharge transient are as follows: 1. The discharge transient current produced at the terminals of the transistor, diode, or integrated circuit must be equal to or exceed IF/D, where IF is obtained from Equation (8.9). D is a derating factor to account for statistical variations in device failure thresholds. D is 3 for Pp values obtained using measured damage constants, and 10 for Pp values obtained using damage constants from generic data tables or those obtained from analytical expressions.

294

8. Space Environment and Interference

2. The discharge transient voltage produced at the t e ~ i n a l s of a transistor, diode, or integrated circuit must be equal to or exceed VvO, where VFD = V B + 1FR~/D.

References 1.

Abraham L. Pressman, "Switching Power Supply Design," McGraw-Hill, New York.

2.

Y.S. Lee and Y. C. Cheng, "Computer Aided Analysis of Electronic DC-DC Converters," tEEE Trans. Aerospace Electronics Systems 24(12) 124-132, March 1988.

3..

E. T: Moore and T. G. Wilson, "Basic consideration for DC to DC conversion network.," IEEE Trans. Magnetics 2(3), September 1993.

4.

Lazar Rozennblat, "Understanding the transient response of switching mode supply, .... Electronic Design, November 1997.

5.

N.E. Lindebland, U.So Patent 2, 300,052 filed May 4, 1940 issued October 27, I942.

6.

D. Rubin, "The Linville method of high frequency transistor amplifiers design," Naval Weapons Center, Research Department, NWCCL TP 845, Corona Laboratories, Corona, CA., March 1969.

7.

S.W. Lee and Yr.Rahmat-Samii, "Simple formulas for desiring an offset multibeam parabolic reflector," IEEE Transactions in. Antennas Propagation, AP-29(3), p. 472 (May 1981).

8.

L Ruze, "Antenna tolerance theory--a, review," Proc. IEEE, 54, pp. 663-640 (April 1996).

9.

M.i. Kontovitch, "Average boundary conditions at the surface of the grating with square mesh," Radio Engineering (1962). M.I. Astrakhan, "Reflecting and screerfing properties of plane wire grids," Radio Engineering, 23 (1968).

10. 11.

E Dumont and E Combes, "Study of reflection by metallic mesh," Int. Syrup. Santiago de Compostella, Spain, August 23-26, 1983.

1.2.

E Dumont, "Modelisation radioelectrique des antennes a reflecteur deployable," Thes. Doc. Eng. ENSAE, Totouse, France (1984).

t3.

S.W. Lee and Y, Rahmat-Samii, "'Vector diffraction analysis of relector antennas with mesh surfaces," IEEE Transactions in Antennas and Propagation, 33, pp. 76-90 (January i985).

14a. V.A. Davis and L.. W. Duncan, "Spacecraft Surface Charging Handbook," P L ~ 92-2232, Maxwell Laboratories for Philips Laboratory, Air Force Material Command, Nov. 1992. 14b. C. K. Purvis, K. Ga,~et, A. Whittlesey, and N. J. Stevens, "Design guidelines for assessing and controlling spacecraft charging effects," NASA TP2361 (1984),

References

295

I5. D. A. Hardy et al., "Average and worst case specifications of precipitating auroral ei~tron environment," Spacecraft Environmental Interactions Technology 1983, NASA CP~2359, AFGL-TR-85-00!8, ADA 202020, edited by C~ K. Purvis and C. E Pikes, pp. 13I-I53 (1985). 16. C. S. Lin and R. A. Hoffman, '°Characteristics of the Inverted V Event," Journal of Geoplz~4cal Research. ~ , p. 15 ! 4 (1979). 17. C. S. Lin and R. A. Hoffman, "Nucmations if lnve~ed V Fluxes," Journal of Geoph3~ics Research, ~ , p. 6471 (1979). 18. E. G. Multen and M. S. Gussenhoven, ~'High-Leve! Spacecraft Charging Environments Near Geosynchronous Orbit,"AFGL-TR-82-0063, ADA 118791 (1982). 19. M. S. Gussenhoven et al, '~High, level spacecraft charging in the low-altitude polar auroral environments," Journal of Geophysical Research. 90, pp. 11009-.11023 (t985). 20. E. G. Fontheim et at., "S~tistical study of precipitating electron~" Journal of. Geophysical Research, 87(A5), pp~ 3469-3480 (1982)~ 21. H. C. Koons et al., "Severe spacecraft charging event on SCATHA in September t982," Journal of Spacecraft and Rockets, 25, pp. 239-243 (1988). 22. H. C. Koons and D. J. Gorney, "Relationship between electrostatic discharges on spacecratt P78~2 and the electron environment," Journal of Slmcecraft and Rockets, 28(6), pp. 683-688 (199t). 23. H, B. Garret et at., "Environment-induced anomalies on the TDRS and the role of spacecraft charging," AIAA paper 90-0178, 1990. 24. I. Katz et at., "The importance of accurate seconda~ electron yields in modeling spacecraft charging," Journal q[ Geophysical Research, 91, pp. t 3739..--13744 (1986). 25. U. Fahleson, "Plasma vehicle interactions in space--some aspects in present knowledge and future development," Photon and Partic& Interactions with Surfaces in Space. edited by R. J. L. Grard, D. Reidel Publishing Co., Dordrecht, Holland, pp. 563-569, 1960. 26. M. J. Mandetl, NASCAP Programmer's Reference Manual, S-Cubed Division, La Jotla, CA SSS-R-84-6638, 1984. 27. K. G. Balmain, "Thickness scaling for arc dischalges on electron beam charged dielectrics," IEEE Trans. on Nuclear Sciences, NS-32(6), pp. 4073-4078 (1985). 28. J. Wilkenfeld et ai., "Development of Electrical Test Procedures for Qualification of Spacecraft against EID," Vol. t, IRT 8195-018, IRT Corp. 1981. 29. E.W. Grey, "Vacuum surfaces Jlashover: a high pressure phenomenon," Journal ~ Applied Pkysics, 58(I ), p. 32 (1985). 30. R. Stetmer and A. B. Dewald, "A surface discharge model tbr spacecraft dielectrics," IEEE Trans. on Nuclear Science, NS-32(6), pp. 4079--4086 (!985).

296

8. Space Environment and. Interference

31. A. R. Krauss, "Localized plasma sheath model on dielectric discharge of spacecraft polymers," IEEE Trans. on Nuclear Science, NS-35(6)(1988). 32. C. Bowman et al., "Space-charge level current injection testing to investigate discharge coupling models," IEEE Trans. on Nuclear Science, ~(6), p. 2033 (1989). 33. L E Granger and J. G. Ferrante, "Electrostatic discharge coupling in spacecraft electronics," ESA Journal, 11(14), pp. 19-30 (1987). 34. G. J. Burke and A. J. Poggio, Numerical Electromagnetic Code (NEC), Part 1 and Part 22: Program description, Lawrence Livermore National Lab. UCID-18834, 1981. 35. A. J. Woods et al, "Model of coupling of discharges into spac~raft structures," Spacecraft Charging Technology-- 1980, NASA CP-2182, AFGL-~-81 ~0270, ADA 114426, ~ited by N. J. Stevens, pp. 745-754 (1981). 36. A. J. Woods and E. E Wenaas, "Spacecraft discharge electromagnetic interference coupling models," Journal of Spacecraft and Rockets, 22(3), pp. 265--281 (1985). 37. J. E Granger and J. G. Ferrante, "Electrostatic discharge coupling in spacecraft electronics," ESA Journal, II(14), pp. 19-30 (1987).

Index Note: Page numbers in. italics refer to the figure or table on that page.

Absolute charging, 261 Acceleration of launch vehicle, 8 Accelemmeters, 78-80 Amplitudeoshift keying (ASK). 160 modulation, 162 Anomalies, 45, 47 Antenna factor. 229-.230 Antennas elements, 233-234 steering, 23 theory, 222-229 types, 234 Antenna systems, 233-.242 Apogee, Arcing, 280-282 Ariane launch family, i0-11 Articulation control for satellites, 2 Ascending node.. 45 Assembly of launch vehicles, 14~t 5 At!as-Centaur, 14~ 15. Attitude and articu!ation control subsystem (AACS),. 31, 48 earth sensors, 49 interference problems. 31 ~ 7 Attitude control, 2, 48~96 thrusters, 57 Auroral environment, 255-2.58

Bandwidth communication for digital signals, 160-17/ Batteries, 134~13.9 charging, 137-I 39 NiH z, I36-137

Beam4brming networks, 239-24I Besbcase analysis, 180-181 Binary coding, I55-I57 Binary-phase shift keying (BPSK), 161, I63 Bipolar signal, 156 Blowoff discharges, 281-282 Boost converter, Itl Buck converter, 1 ~ 1 0 7 Bypass capacitors., 8%92 in DC-DC converters, 131

Cable shields, 95, 96 Cadence WCA tool, 172 Cassegrain antennas, 236 Celestial star assembly (CSA), 49 Charged coupled device (CCD) in star camera, 65 Charging auroral environment, 255-258 auroral vs. geosynchronous, 263 currents, 268-274 effect of electron energy, 25%260 effects of, 261-263 environments, 266-268, 269 in geosynchronous environment, 252-255, 26 I multibody interactions, 261 satellite surface, 260~264 Charging potential calculatiom 257 Circuit failure, 293-294 Circuit upset, 28~288 Circular orbits, 40 Clock waveforms, 6%70 Code-division multiple access (CDMA), 21-22, 212~ 217-221 Globalstar system, 2I Odyssey system, 27 297

298

Index

Coding, 148, 155-171 Command and data handling, 3-4. t 4 I~ 148 requirements, 142~143 subsystem (C&DH), 49 Command interlace control unit (C1CU). 1 ~ Commercial launch vehicles, 12-I 5 Common-mode current, 72-73 Common-mode rejection, 9 t-92 Component damage, 28%294 Conducted interference in SMPS, I 13-I i4 Converters coupled to transformers, 10% 109 in SMPS, I26 types, I05~ 1 Coriolis acceleration, 32 Countries with samllites, 13 Coupled cavity TWTA, 205 Coupling analysis, 284-286 betw~n antennas, 23~232 between samllites, ! 7 Crosstalk between satellites, I7 Cuk converter, 107, 1!3 Current collection from a plasma. 2 ~ 2 6 6 common- and differential-mode, 72-73 limiting inrush, 112-113 in SMPS, 122, 125 substorm vs. aurora, 272-274 Current noise, 83

Damage to circuits, 289-294 to components, 28%294 m integrated circuits, 291 Data handling for satellites, 3 4 DC-DC converters, 126-.127 ground and power layouts, 132 optimized design, 135 Decoupling, 179 Decoupling of suNractors, 93 Delta pulse code modulation (DPCM), 158-1 Descending node, 45 Design control of spacecraft charging, 263 laser ine~iat measuremer~t unit, 7%78 reaction wheels, 57"58 reflector antennas, 244 satellite communication systems, 5~8

solar arrays, 102-103 sun sensors, 51-52 Device capacitive loading, 175 Device electrical characteristics, t 82-I 84 Device loading factors, 18 !-182 Dielectric breakdown, 281 Differential charging, 261,275.-280 effect, 261-262 shade~unshaded surface, 27~279 Difl?rentiaPmode current, 72-73 Differential potential of sides of spacecraft, 27%280 Digital modulation, 6-7, !4%153 Digital signaling methods, 160-t 71 comparison, 170 Diode damage, 292 function, l t 5-116 switched OFF operation, 120 switched ON operation, 1t9 Discharge energy path, 282-286 Discharging, 280~.282 Distortions in TW~FAs, 208~2t I Dual-reflector antennas, 236

Earth stations. 7 Eastern Test Range, 10, 12 ~lipse number for LEO,. 134, 135 Electrical connections, 5 Electric field calculation, 22%230 Electron current calculation, 270 Elliptical motion, 35-36 Elliptical orbits, 40-4 I EMC Radiative couNing method, 284 EMI noise, 120-I 21 avoiding, 8%92 from diodes, I t 5-116 filter in SMPS, 126-127 Energy storage, 134-139 Engineering telemetry unit (ETU), 148 Equilibrium potential contours, 276

Feed systems, 23%242 Field programmable gate array (FPGA), 77 Field regions of antennas, 222 Flight processor computer~ I43-144

Index. Floating potential, 264 Flyback converter, It6 Fontheim distributions, 25%258 Forward converter, 107-109 with parasitic elements, 124 Fraunhofer region, 222 Frequency-division multiple access (FDMA), 2t2, 2t3 Frequency reuse, 24%.250 Frequency~shift keying (FSK), 1,61, I64-!65 Fresnel region, 222

Gain., 209 Gate.ways, 7 Geosynchmnous environment, 252-255 Geosynchmnous orbit (GEO), 15, 17-18 Geosynchmnous plasma, 253-255 Gimbal drive electronics (GDE), I46-147 Globalst~ system, 18, 2 t-22 Granular noise, 155 Gray code, 151 Gregorian antennas, 236 Grounding to avoid EMI noise, 89-92 op-amps, 92,-96 SMPS convermrs in PCBs, 125-133 sun sensors, 54-55 to suppress transient effects in SMPS, !2%131 Guard shields, 95 Guidance subsystem. See Attitude and articulation control subsystem Gyroscopes, 74-77 in laser inertial measurement units, 77

Harness, 5 Helix TWTA, 205-206 Horizon sensors, 49 Hunting noise, I55 Hyperbolic motion, 36 Hyperbolic orbits, 4!

Impedance of an antenna, 225-226 Impulse calculation, 8 Inertial measurement unit (tMU), 49, 74-96 laser, 77-78

299

Input noise of power converiers, 110 Input/output board WCA, 184-190 Input ripple noise of power converters, 111 Integrated circuit damage, 292 Interference coupling ~tween antennas, 230-232 Interference paths in SMPS, 122, I25 Intermodulation distortion, 2 I(3-21 t lntersaiellite links for Teledesic system, 26 !ntraboard and connector capacitance, 176 Ion current calculation, 270 Iridium satellite system, 20-2I

Johnson noise, 81, 86

Kep!erian orbits, 38*-4I Kepler's laws, 39 Klystron, 205

Langmuir probe, 264 Laser inertial measurement unit, 77-78 Launch sites, 1O, 12 Launch vehicles, 8-11 assembly, 14-15 commercial, 12.~,,[5 man ufacturers, 13- ! 4 for personal wireless communication, 12-15 Lens antennas, 235 Line delay correction, 178 Line of apsides, 4 4 4 5 Lines of n~es, 45 L~kheed-Martin wrap-rib antenna, 245-249 Ia3w~earth orbits (LEO), 18-20 Lumped element method (LEM), 282, 284-.286

M Magnetosphere, 251-252 Manchester signal, 156 Manufacturers of launch vehicles, t 3-i 4 Masmr crystal oscillator (MXO), 146 Mechanical gyroscope~ 74~75 Mechanical systems in satellites, 4 Mentor WCA too!, I72

300

Index

Mobile satellite communications, 15~30 Mobile satellite multichannel requirements, 249-250 Modulation, 6 Modulation theory, t48-17 ! Motors emf calculation, 60 in reaction wheels, 56 torque calculation, 60 Multibeam antenna system, 241~242 Multilevel. signaling, 165~I71 Multiple access, 212-.2:2I comparison of types, 219 Multiple access system, 7 Multiplexing, 6

N NiH 2 battery, 136-.137 Noise in analog~to-digital converters, 8 ~ 8 2 electromagnetic coupling, 7(t-72 frequency characteristics in oDamps, 84 in high speed sampling of A/D converters, 82-84 intrinsic in op-amps. 63-64 PCB Layout, 130-13 t in power electronics, !09-I 40 from power transistors, 120-122 in pulse code modulation, 153-t55 quantizing, t 55 in reaction wheel assembly, 58~62 from silicon-controlled rectifiers, l 16-120 sources, 69-70 in st~ camera, 66-69 in sun sensor circuits, 52-55 in switching-mode ~ w e r supplies., 10% 133 in telecommunications subsystems, 193-201 temperatum~ 199 tomi input, 85-87 Noise figure, I99,-200 Numerical electromagnetic method, 283

Odyssey system, 18 Offset-fed parabolic antenna design curves, 238 design formulas, 240 Op-amps grounding, 92-96

intrinsic noise. 63-64 noise figure, 84 power supply decoupling, 8%92 total noise outpuL 85~87 Operating environment device derating, 180 Orbitdimited current collection, 264-266 Orbits circular, 40 configuratiol~ vs. satellite coverage, 43 defining, 45 determination, 33~38 elliptical 40-4 I hype.rbolic, 41 Keplerian, 38M ! I.~ation and power system design, 10l parabolic. 4 I types and attitude control, 48 Orthogonal access techniques, 212 Oscillation in sun sensors, 53~.-64 Overload noise. 155

Packet switching technology, 24 Parabolic motion, 37-38 Parabolic orbits, 4t Parasitic currents, 73 Parasitic extraction tools, t3!-I33 Particle-pushing method, 283.-284 Path: of discharge energy, 282-286 Payload data. assembly (PDA), 148 Payloads, 4-5 PCB. See Printed circuit board Pendulous gym inwgrating accelerometer (PGIA), 7%80 Perifoca!, 44 Perigee, 44 Periheli(m, Phase array antennas, 234-235 Physics of rocket motion, 8-9 PN junction, 98 Polar signals~ 155 Potential, rate of change. 276 Power average radiamd by an anwnn.a, 223 calculations for antennas, 22~227, 228 calculations for solar cell, 99 control electronics, 137M39 gain of an antenna, 224 output vs. input, 2t 1 received ~ w e r calculation, t 96-- ! 98 Power converters, 105.-~109

Index Power gain of antennas, 222 Power source for satellites, t-2 Power spectral density binary' code signals, t56~157 binap¢ phase-shift keying, t63 digital bandpass signak I60 frequency~shift keying, 164-165 quadrature amplitude modulation, ~67, 168 Power subsystems, 97-I08 Power supply decoupling in op-amps, 8%92 rejection ratio (PSRR), 87, 89 Printed circuit board layout and ndse reduction, 130-I 3 I layout using parasitic extraction tools, 1.31-133 Propagation delay, 173-~174 correction factor, 176-t 78 end-of-life correction, t,78 open collector calculation., 177-I 78 Propulsion control in satellites, 4 Public switching telephone network (PSTN), 5 Pulse amplitude modulation (PAM), 150M 51 Push-pull converter, 109, ll 7 Pyrotechnic relay assembly, 147

QPSK modulation, 20 Quadraphase shift keying (QPSK), 165~t66 Quadrature amplitude modulation (QAM), I67~i7I

Radial-rib antenna, 245 Radiation damage m solar cells, 103~104 Radiation pattern of antennas, 222 Random noise., 155 Reaction wheel assembly, 49 Reaction wheels, 55-62 design.,. 5Z~58 functions, 55 Received power calculations, 196-t 98 Refere..nce bursts, 2 ! 6 ReNector antennas, 235-238 Ring laser gyroscope, 75~.77 RMS, 63 output noise, 86 Rockets, See Launch vehicles Routing priority lbr Odyssey system, 28

301

Sagnac effect, 75-76 Samllite charging effects, 260-264 communications, 5--8 covemge~ 41-44 defining orbital ~sition, 45, 47 development, 15 as floating pr..o~s, 264~266 mobile communications, 15-30 navigation, 31 space between, 17 subsystems, 1-5 Satellite systems Globalstar, 18, 21-22 Iridium, 20-2 I Odyssey, t 8, 2%30 store and dump, 18 Tetedesic, 22-27 SCATHA satellite, 254, 261 Semiconductor damage, 2.90 constants, 291 Sensors horizons 64 star, ~ 6 6 sun, 50-54 Set-up and hold sr~cificaaons, 178-I 79 Shields in op-amps, 92-96 Signal, 5=8, base band, 14 l carrier, 14 l formats for binary coding, 155 gain, 209 modulation, 6 processing, 5-8 processing by command system, 141- ! 48 Signal control unit (SCU), 145-i46 SignaMo-noise ratio delta modulation system, !~Ki pulse code modulation, I54 Silicon-controlled rectifiers, ! 16-I 20 Single-reflector antenna, 236 Solar arrays, t01-103 So~ar cells, 98~i radiation damage m, 103-!04 in space systems, 10~102 Solar energy, 97 Solar flares, 1 Solid state power amplifiers (SSPA), 202 Sources of interference in SMPS, i 14~-122

302

Index

Sources of l~oise in A/D converters, 81 model, 85-87 in opoamps, 82 pulse code modulation, 153 in TWTA, 207-208 Space-charge limited current collection, 264-266 Spacecraft. See Satellites Spacing for satellites, I7 SPICE-like WCA tools, 173 Spin-stabilized satelliws, 48~49 Spread-spectrum effect, 217 Star camera, ~ 6 6 optical system, 65-66 Steering an.mnna, 23 Stettner-DeWald modeI, 281 Storage cell selection, I35 Strip-aperture model unfurlable antenna, 247 Structure of satellites, 1 Sunlight/eclipse ratio, t01 Sun sensors, 49 design, 51-52 grounding, 54-55 physical Ninciples, 50~55 types, 50 Surface charging,. 262-263 Su.r~ace coating and differential charging, 275-280 Switching Iridium satellite system, 21 transistors, 62 Switching-mode power supply (SMPS), 105-109 current, 125 design, i26 interference paths, 122, ! 25 transient effects in, 12%130

Traffic burst archflecture, 216 ~?affic management~ See Switching Transistor damage, 292 Transistor switching process, 62 Transition times, 18 l ~lS:ansponder noise representations, 202-221 Traveling wave tube amplifiers (TWTAs), 202, 205-208 Two-body central force motion, 31-33 Types of launch vehicles, 10-t t of signals, 5

Unfu.rlable antennas, 245-249 e×ample~ 245-~249 Unipolar signal, 155 Uplink/downlink equations, 200-201. UplinkJdownlink models, 194- t 96 Uplink pwcessor WCA, 190-I 93

V~: variation, I75 "Velocity of rocket ascent, 8 Vis-viva equation, 39-40 Voltage and carrent waveforms boost converter, 11.2 buck-boost converter, 110 buck convener, I08 flyback converter, 117 forward converter, t15 push-pull converter, i18 transistor, t22 "Voltage noise of op-amps, 83 W

Tel~esic satellite system, 22-~27 Temperature balance in satellites, 4 Thermal control for satellites, 4 The~at runaway, 290 Thrusters, 57 Time-division multiple access (TDMA), 20, 212, 214~216 architecture,. 215-216 Titan launch family, t0-1 I Tools l~r worst-case analysis, 172-173 Torque calculation mechanical gyroscopes, 75 motors, 60

WCA. See Worst-case analysis Western Test Range, t 0, 12 Wire-grid mesh unfurlable antenna, 246-247 Worst~case analysis digital circuits, 173M79 examples, 17%193 guidelines, 17 l-172 power electronics, t 3%140 tools, !72-173 Worst case charging events, 254, 257