IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 52, NO. 8, AUGUST 2004
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Compact Wide-Band Multimode Antennas for MIMO and Diversity Christian Waldschmidt, Student Member, IEEE, and Werner Wiesbeck, Fellow, IEEE
Abstract—This paper presents broadband multimode antennas for multiple-input multiple-output (MIMO) and diversity applications. The antenna system is not based on spatial diversity, as usual MIMO systems, but on a combination of pattern and polarization diversity. Different modes of self-complementary, thus extremely broadband, spiral and sinuous antennas are used to decorrelate the signals. It is shown that only one antenna is necessary to receive three uncorrelated signals, thus the space required to place the MIMO antenna is very small. Simulation results and measurements of a typical indoor scenario are given. Index Terms—Multimode diversity, multiple-input multipleoutput (MIMO), sinuous antenna, spiral antenna.
I. INTRODUCTION
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UTURE communication systems have to fulfill the requirements of high data rates and flexible interfaces for different communication system standards. Multistandard radios, offering the demanded flexibility to use different standards, require very broadband antennas. multiple-input multiple-output (MIMO) and diversity systems allow exploitation of the spatial channel properties. If the signals received by different antennas are uncorrelated, very high data rates may be reached as recent studies have shown, first in [1] and later in [2], [3]. Usually uncorrelated signals are obtained by spatial diversity, which requires large antenna spacings. This paper presents new broadband antenna solutions, that are small enough to fit into laptops or organizers, but that still yield uncorrelated signals for MIMO or diversity applications. The compactness of the broadband MIMO antenna system is not achieved by using different antennas, but by one antenna with different, independently fed, modes. This results in multimode diversity, a combination of pattern- and polarization diversity to obtain uncorrelated channel impulse responses for the MIMO or diversity system. As far as the authors are aware multimode diversity has first been suggested in [4], where orthogonal azimuth patterns were used. In [5] a multimode patch antenna with different modes for diversity was presented. Multimode diversity for MIMO has been suggested in [6], but this paper presents a new and practical antenna concept, based on spiral and sinuous antennas. In [7] the ability of logarithmic spiral antennas to radiate in different polarizations is discussed and a possible application for diversity is mentioned, but not explicated. Besides uncorrelated signals at the antennas, which are obtained by orthogonal patterns the mean signal to noise ratio Manuscript received February 4, 2003; revised August 25, 2003. The authors are with the Institut für Höchstfrequenztechnik und Elektronik (IHE), Universität Karlsruhe (TH), Karlsruhe D-76128 Germany (e-mail:
[email protected]). Digital Object Identifier 10.1109/TAP.2004.832495
Fig. 1. Geometry of a spiral antenna with voltage sources between the single arms of the spiral.
(SNR) of all the signals has to be “similar,” see [8], to obtain a diversity gain or capable MIMO systems. “Similar” in this context means, e.g., less than 10 dB difference for two branch maximum ratio combining, [8]. In this paper it is shown, that the mean effective gain (MEG), which is linked to the SNR, of the single modes differs by only 1 to 2 dB, thus a high diversity gain is obtained. For MIMO the total received power or the mean SNR respectively is an important quality measure for an antenna array. By a comparison with a dipole array with large antenna spacings, which is generally considered as a capable array for MIMO, the ability of multimode antennas for MIMO is shown. This paper is organized as follows. In the first section four-arm spiral and sinuous antennas and the different excitations for the modes are presented. Second, the correlation properties of signals received by different modes of the antenna and the mean effective gains are given as a function of the incident field and its spatial distribution. In the last section MIMO capacity calculations and measurements with spiral antennas are given. II. SPIRAL AND SINUOUS ANTENNAS The self-complementary, archimedian, four-arm spiral antenna and sinuous antennas are well described in the literature, see, e.g., [9]–[11], thus only the properties crucial for multimode diversity are given here. The spiral antenna consists of four arms, that are rotated around the center of the antenna, see Fig. 1. The antenna can basically radiate three different modes depending on the excitation. For this application mode 1 and mode 2 are used. Mode 1 is characterized by a phase shift of 90 between adjacent sources at the single arms of the spiral, see Fig. 1. Mode 2 has a phase shift of 180 . Both modes are circularly polarized in the direction of the main radiation
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Fig. 2. Geometry of a four-arm sinuous antenna. describes the lengths of the teeth and is therefore a determining antenna parameter.
Fig. 4. Pattern of mode 2 of the spiral antenna with a radius of 10 cm at 2 GHz. The pattern hardly changes versus frequency for frequencies above 1.2 GHz.
Fig. 3. Pattern of mode 1 of the spiral antenna with a radius of 10 cm at 2 GHz separated into left (lhc) and right hand circular (rhc) polarization. If the spiral is fed at the outer end of the arms, the polarization is orthogonal to the one obtained by exciting at the center of the spiral.
and elliptically polarized otherwise. Due to the self-complementarily the antennas are frequency-independent or, in other words, extremely broadband. Since the geometrical structure of the spiral antenna is finite, there exists a lower frequency bound. This bound is (1) is the speed of light, the outer radius of the spiral where the effective substrate permittivity. has to be deand termined by simulations of the spiral antenna or experimentally. is close to one for etched spirals, According to experience, also for a high of the substrate. For all simulations presented in this paper the antennas were simulated with FEKO [12], a standard software tool based on method of moments. Equation (1) is explicable by the current distribution within the active zones of the single modes [10]. The active zone is a circular area located around the center of the antenna. The energy is radiated from the antenna in the active zone. This zone is characterized by a certain ratio of its circumference to the wavelength. For
Fig. 5. Phase of the pattern of mode 1 and mode 2, shown in Figs. 3 and 4. The phase of mode 1 changes 360 per circulation around the antenna, mode 2 changes 720 .
mode 1 the circumference is one wavelength, for mode 2 it is two wavelengths. Thus, the current distribution on the arms of the spiral in the active zone has two maxima for mode 1 and four for mode 2. Above this lower frequency bound all antenna properties are almost stable and change only slightly with freof mode 1 and 2 are given quency. The pattern in elevation in Fig. 3 and Fig. 4. The azimuth patterns are omnidirectional. The phase of the complex radiation pattern, which among other parameters determines the correlation among the receive signals, is shown in Fig. 5. The phase of mode 1 changes 360 and the one of mode 2 720 for each circulation around the antenna, which is explicable by the current distribution within the active zones. The modes can be excited in two ways: first by feeding the spiral arms at the inner ends that is at the center of the spiral and second at the outer ends of the arms. Those modes are orthogonally polarized left-hand circular (lhc) and right-hand circular (rhc). The third mode of the spiral antenna (270 phase shift between adjacent arms at the excitation) has a pattern, whose amplitude is equal to mode 1, but the polarization changes from lhc to rhc. Thu,s mode 1 and mode 3 are orthogonally polarized. The unwrapped phase of the pattern of mode 3 changes 1080 per circulation around the antenna.
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Fig. 6. Pattern of mode 2 of the sinuous antenna with a radius of 10 cm at 2 GHz. The pattern hardly changes versus frequency for frequencies above 1.7 GHz. Fig. 8. Gain of the rhc and lhc polarized field of the sinuous antenna with = 50 and an outer radius of 0.1 m.
of the pattern of mode 3 is equivalent to mode 1, but both modes are orthogonally polarized. III. MULTIMODE-DIVERSITY
Fig. 7. Pattern of mode 1 of the sinuous antenna with a radius of 10 cm at 2 GHz. is 50 . Mode 3 is orthogonally polarized, but has the same pattern.
The geometry of a four-arm sinuous antenna is given in Fig. 2 and described in detail in [11]. The antenna is self-complementary and used as a multimode antenna. The modes are excited the same way as for the spiral antenna. The lower frequency bounds of the modes are a function of different geometry parameters, thus are not as easy accessible as for the spiral. In general the lower frequency bounds are higher than the ones for the spiral antenna for a given outer radius of the antennas. They decrease with increasing (for see Fig. 2), since the antenna resembles in sections a spiral antenna for large . The patterns of mode 1 and 2 are given in Figs. 6 and 7. The shapes of the patterns change only slightly with frequency or , but the polarization changes. The pattern is alternately left and right hand elliptically polarized versus frequency, see Fig. 8. The axial ratio of the sinuous antenna depends on . For large the antenna acts in sections like a spiral, thus the axial ratio is almost 0 dB. For small the antenna is rather linearly polarized. Both modes may be excited at the center or at the outer ends of the arms, but in contrast to the spiral antenna, orthogonal polarizations are only , spiral-like behavior). The shape obtained for large (
MIMO transmission channels are characterized by the channel matrix , which contains the channel impulse responses or the channel coefficient in the flat fading case between the different sets of transmit and receive antenna ports. For broadband systems the spectrum can be divided into narrowband sections with flat fading. The diversity gain or MIMO capacity depends on the correlation coefficients among those channel coefficients of , see [3], and the SNR. The correlation is influenced by the statistical properties of the wave propagation and the antenna properties, in this case the properties of the single modes. In the following the correlation coefficient among two receive signals as a function of the incident field is calculated. This is equivalent to the correlation for one transmit and two among the channel coefficients of receive antennas in a MIMO system. The spatial wave propagation properties are describable by and the power azimuth and elevation profile for both polarizations and . To allow for analytical calculations typical statistical functions to model the wave propagation are chosen. Measurements have shown, that the power azimuth is best modeled by a Laplacian function [13] spectrum a for both polarizations. For the power elevation profile Gaussian function is assumed. The total power angle spectrum is given by the product of the Laplacian function for the azimuth and a Gaussian function for the elevation, normalized so that . With [14] (earlier shown in [8] in a similar way) it can easily be shown that the complex correlation coefficient among two signals received by different antennas, in this case different modes, is given by (2)
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Fig. 9. Correlation coefficient among mode 1 (a) and mode 2 (c), excited at the center of the spiral, and mode 1 (b), excited at the outer edge of the spiral to generate orthogonal polarizations. The incident field has an elevation spread of 5 and azimuth spread of 60 . The lower frequency bound of mode 2 is 1.2 GHz, thus the spiral does not work correctly for lower frequencies.
Fig. 10. Correlation coefficient among the same modes as in Fig. 9, but the incident field has an elevation spread of 5 and an azimuth spread of 20 . Due to slight changes in the pattern for different frequencies the correlation coefficient changes. But it is over the whole frequency range low enough to obtain a diversity gain.
with the covariance
(3) where
is constant and the variance
(4) is the ratio of the power in -polarization to the where is a funcpower in polarization at the receiver. Note, that tion of the polarimetric radiation pattern, thus disappears for orthogonally polarized antennas in this case lhc and rhc polarized modes. The power correlation coefficient is obtained by , according to [15]. Basically it is possible to use spiral or sinuous antennas with any different modes and polarizations for multimode-diversity. In the following first a spiral and second a sinuous antenna are used to calculate the correlation coefficients among receive signals. The orientation of the both antenna planes is vertical. A spiral antenna with mode 1 and mode 2, excited at the center of the antenna, and a third mode (mode 1) with orthogonal polarization excited at the outer edge of the antenna is used. Figs. 9 between and 10 show the power correlation coefficient different modes for a large azimuth angular spread of 60 and a small spread of 20 for a mean direction of 10 in azimuth and 0 in elevation of the incident waves. The third mode is orthogonally polarized to the other modes, thus the correlation is almost zero. The other modes are more strongly correlated as the pattern of mode 1 and 2 partly overlap. On the other hand,
Fig. 11. Correlation coefficient among mode 1 (1), mode 2 (2) and mode 3 (3) of the sinuous antenna. The incident field has an elevation spread of 5 and azimuth spread of 60 . The lower frequency bound of mode 2 is 1.7 GHz, thus the antenna does not work correctly for lower frequencies.
the different phases of the patterns of mode 1 and 2 (see Fig. 5) decorrelate the received signals, since the single plane waves from different directions superpose differently for each mode. The influence of the feed network on the pattern of the modes is neglected. is used with three modes, all The sinuous antenna excited at the center of the antenna. Figs. 11 and 12 show the correlation coefficient versus frequency for the scenarios mentioned above. Mode 3 is orthogonally polarized to mode 1 and 2, thus the correlation is low. Mode 1 and 2 hardly overlap, thus different signals are received. In order to fulfill the requirement of an equal or “similar” SNR of the signals received by different modes to obtain a diversity gain the MEG may be used, see [16]. The MEG is defined as the ratio of the mean received power of one antenna under test to the mean received power of a reference antenna, when both
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A. Simulations of the Capacity
Fig. 12. Correlation coefficient among the same modes as in Fig. 11, but the incident field has an elevation spread of 5 and an azimuth spread of 20 . With decreasing angular spread the correlation increases. TABLE I MEG OF DIFFERENT MODES IN DECIBELS (ELEVATION ANGULAR SPREAD 5 )
The channel model used to calculate the capacity of MIMO systems consisting of one spiral antenna at the transmitter and one at the receiver is an extended version of the model described in [19]. This stochastic channel model is based on ray-tracing simulations and measurement campaigns in indoor scenarios. It is a three dimensional double-directional channel model, in other words provides the angle of departure and arrival of each path. The channel model takes only non line-of-sight (NLOS) connections into account. The power azimuth spectrum is modeled by multiple Laplacian functions, each modeling a cluster of scatterers. The elevation profile is modeled by a sine function. The cross polarization coupling is 8 dB. The antennas used for the simulation are one spiral antenna at the transmitter and one at the receiver. Mode 1 and 2 are excited at the center of the spiral, and mode 1 with orthogonal polarization is excited at the outer edge of the spiral. Thus, the same modes as in Section III at both transmitter and receiver are used. The orientation of the spiral plane is again vertical. The result of the simulations with this channel model are channel matrices (obtained the same way as in [20]). Therefore, the capacity of a MIMO system with no channel state information at the transmitter can be calculated [2] SNR
antennas are used in the same channel with the same transmit antenna, see [17], [18]. For the assumptions on the wave propagation made above the MEG can be calculated analytically for an isotropic reference antenna.
(5) are the gain patterns for both polarizations. Table I where shows the MEGs for different antennas and modes for a crosspolarization coupling of 8 dB. The MEGs of mode 3 of both antennas are equal to the one of mode 1. Since the requirement of orthogonal patterns, i.e., uncorrelated signals, and similar MEGs are fulfilled, a diversity gain over a large bandwidth with both antenna types, used as multimode antennas, is obtained.
(6)
conjugate complex transpose and where SNR denotes SNR is the number of transmit antennas, in this case the number of different modes. The channel matrices in (6) are normalized with (7)
to obtain a constant mean gain of each channel matrix, see [21]. The SNR in (7) is the average SNR at the receiver. This normalization allows to show the influence on the capacity of the correlation properties and the distribution of the mean gains of the channel coefficients. This distribution influences the capacity. The channel coefficients between co-polarized modes have a larger mean gain than those between cross-polarized modes. Thus, the mean gains of the channel coefficients are not equal. Equality is considered to be optimal, [2]. Fig. 14 shows the capacity distribution for a constant mean SNR at the receiver of 10 dB for 1000 channel realizations at 2 GHz. The 10% outage capacity is approximately 7.3 bit/s/Hz.
IV. MIMO SYSTEMS BASED ON MULTIMODE-DIVERSITY In order to show the potential of multimode antennas in MIMO systems, simulations and measurements of the capacity of a MIMO system with one multimode spiral antenna on each side of the link, were performed. Additionally a comparison with dipole antennas, arranged in parallel, is drawn. For the simulations a sophisticated channel model is used. This model does not allow for analytical calculations like in Section III, but it allows to assess the MIMO performance in very realistic environments.
B. Power Considerations When the normalization in (7) is used in other words when the gain of each channel matrix is normalized, the information about it is lost. But to assess arrays for MIMO completely, this information needs to be taken into account to assure a high efficiency of the complete MIMO channel. Fig. 15 shows the cumulative distribution function of the gain of the channel matrices of the simulations. The comparison with a MIMO system with arrays consisting of three vertical half-wavelength
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Fig. 13. Scenarios for the measurements. For the LOS scenarios transmitter and receiver are placed in the same room. For the NLOS scenarios the transmitter is placed in the corridor.
Fig. 15. Transmission gain of different MIMO systems for the path based channel model. The SISO system has one transmit and one receive dipole.
Fig. 14. Measured cumulative distribution functions of the capacity for different antenna scenarios at 2 GHz. The three dipoles have spacings of =4. The capacity of the NLOS measurements reaches higher capacities as in the LOS scenarios for a constant mean SNR of 10 dB.
dipoles (also simulated with FEKO) with spacings on each side of the link shows, that the gain of the channel matrices of the multimode MIMO system is not worse than with the dipole arrays. Additionally this distribution function is given for dipole spacings on each side of the link. arrays with C. Measurements The measurements were performed with two spiral antennas. The antenna were designed for a frequency range from 1.2 GHz up to 2.5 GHz, limited by the feeding network. Mode 1 and 2 are excited with the feeding network given in [9]. At the outer ends of the arms a hybrid mode with orthogonal polarization compared to the other modes is excited. The coupling between the single modes is below 20 dB. The measurement system consists of a two channel network analyzer, amplifiers and coaxial switches. The channel coefficients were measured one by one. All measurements were done during night, in order to reduce the time variance of the channel. The measurements were performed in an office building, with concrete ceilings and concrete and wood covered walls. The average office size is 4 5 m, see Fig. 13. The receive antennas were placed at the same position
for all measurements. The transmitter was moved along two different routes, shown in Fig. 13. During the first route a strong LOS component is present, whereas the other route is always NLOS. Along each route measurements at 801 discrete frequencies in the frequency range from 1.5 to 2.5 GHz at 210 different positions were performed. The measured data are normalized, according to (7), to obtain a constant mean SNR of each channel matrix of 10 dB. Fig. 14 shows the cumulative capacity distribution for both routes at 2 GHz. The capacity distribution changes negligibly with frequency. Due to the higher multipath richness of the NLOS route, it outperforms the LOS route. For comparison two dipole arrays, consisting of three dipoles each, were used, one at each side of the link. The dipoles were and vertical polarizaarranged in parallel with spacings of tion. The array covers approximately the same area as the spiral antenna with dimensions, so that the resonance frequency of the dipoles equals the lower frequency bound of the spiral. Fig. 14 shows, that the dipoles perform worse than the spiral, since neither polarization nor pattern diversity is exploited. The space diversity is very limited due to the small antenna spacings. V. CONCLUSION This paper shows that four-arm spiral and sinuous antennas allow to exploit multimode diversity, which is a combination of pattern and polarization diversity. The antennas are extremely broadband, thus allow applications for multistandard radios. The space required for the antennas is relatively small. If placing dipoles on the same space required by the spiral, the dipoles do not reach the capacity of multimode-based MIMO-systems. REFERENCES [1] J. H. Winters, “On the capacity of radio communication systems with diversity in a rayleigh fading environment,” IEEE J. Select. Areas Commun., vol. 5, pp. 871–877, May 1987. [2] G. J. Foschini and M. J. Gans, “On limits of wireless communications in a fading environment when using multiple antennas,” Wireless Personal Commun., vol. 6, pp. 311–335, 1998.
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[3] C. Chuah, D. N. C. Tse, J. M. Kahn, and R. A. Valenzuela, “Capacity scaling in MIMO wireless systems under correlated fading,” IEEE Trans. Inform. Theory, vol. 48, pp. 637–650, 2002. [4] E. N. Gilbert, “Energy reception for mobile radio,” Bell Syst. Tech. J., vol. 44, pp. 1779–1803, 1965. [5] R. G. Vaughan and J. B. Andersen, “A multiport patch antenna for mobile communications,” in Proc. 14th Eur. Microwave Conf., 1984, pp. 607–612. [6] T. Svantesson, “An antenna solution for mimo channels: the multimode antenna,” in Conf. Record 34th Asilomar Conf., vol. 2, 2000, pp. 1617–1621. [7] O. K. Kim and J. D. Dyson, “A log-spiral antenna with selectable polarization,” IEEE Trans. Antennas Propagat., vol. AP-19, pp. 675–677, Apr. 1971. [8] R. G. Vaughan and J. B. Andersen, “Antenna diversity in mobile communications,” IEEE Trans. Veh. Technol., vol. 36, no. 4, pp. 149–172, July 1987. [9] E. Gschwendtner and W. Wiesbeck, “Multi-service dual-mode spiral antenna for conformal integration into vehicle roofs,” in Proc. IEEE Int. Symp. Antennas and Propagation AP-S, vol. 3, Davos, Switzerland, 2000, pp. 1532–1535. [10] R. G. Corzine and J. A. Mosko, Four-Arm Spiral Antennas. Norwood, MA: Artech House, 1990. [11] T. T. Chu and H. G. Oltman, “The sinuous antenna,” Microwave Syst., News and Commun. Technol., vol. 18, pp. 40–48, 1988. [12] www.emss.de [Online] [13] K. I. Pedersen, P. M. Mogensen, and B. H. Fleury, “Spatial channel characteristics in outdoor environments and their impact on BS antenna system performance,” in Proc. IEEE Vehicular Technology Conf., 1998, pp. 719–724. [14] K. Fujimoto and J. R. James, Mobile Antenna Systems Handbook. Norwood, MA: Artech House, 1994. [15] J. R. Pierce and S. Stein, “Multiple diversity with nonindependent fading,” in Proce. IRE, vol. 48, 1960, pp. 89–104. [16] M. G. Douglas, M. Okoniewski, and M. A. Stuchly, “Performance of pcs handset antennas in mobile environments,” in Proc. IEEE MTT-S Int. Microwave Symp. Dig., vol. 3, 1997, pp. 1759–1762. [17] J. B. Andersen and F. Hansen, “Antennas for VHF/UHF personal radio: a theoretical and experimental study of characteristics and performance,” IEEE Trans. Veh. Technol., vol. AP-26, pp. 349–357, 1977. [18] T. Taga, “Analysis for mean effective gain in mobile antennas in land mobile radio environments,” IEEE Trans. Veh. Technol., vol. 39, pp. 117–131, 1990. [19] T. Zwick, C. Fischer, and W. Wiesbeck, “A stochastic multipath channel model including path directions for indoor environments,” IEEE J. Select. Areas Commun., vol. 20, pp. 1178–1192, 2002. [20] C. Waldschmidt, T. Fügen, and W. Wiesbeck, “Spiral and dipole antennas for indoor MIMO-systems,” Antennas Wireless Propagat. Lett., vol. 1, no. 9, pp. 176–178, 2002. [21] J. W. Wallace and M. A. Jensen, “Characteristics of measured 4 4 and 10 10 MIMO wireless channel data at 2.4 GHz,” in Proc. IEEE Symp. Antennas and Propagation, vol. 3, 2001, pp. 96–99.
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Christian Waldschmidt (S’01) was born in Basel, Switzerland, in 1976. He received the Dipl.-Ing. (M.S.E.E.) degree in electrical engineering from the Universität Karlsruhe, Karlsruhe, Germany, in 2000, where he is currently working toward the Ph.D. degree. From 2001 to 2003, he was with the Institut für Höchstfrequenztechnik und Elektronik (IHE), Universität Karlsruhe (TH), as a Research Assistant. He serves as a Lecturer for smart antennas and radar antenna systems for the Carl Cranz Series for scientific education. His research activities mainly focus on multiple input multiple output systems, smart antennas, small antennas, integration of antennas and vehicular antennas for radar and mobile communications applications.
Werner Wiesbeck (SM’87–F’94) received the Dipl.Ing. (M.S.E.E.) and the Dr.-Ing. (Ph.D.E.E.) degrees from the Technical University of Munich, Munich, Germany, in 1969 and 1972, respectively. From 1972 to 1983, he was with AEG-Telefunken in various positions including the Head of Research and Development, Microwave Division, Flensburg, Germany, and Marketing Director in the Receiver and Direction Finder Division, Ulm. During this period he had product responsibility for millimeter-wave radars, receivers, direction finders and electronic warfare systems. Since 1983, he has been Director of the Institut für Höchstfrequenztechnik und Elektronik (IHE), University of Karlsruhe, Karlsruhe, Germany, where he is presently Dean of the Faculty of Electrical Engineering. In 1989 and 1994, respectively, he spent a six month sabbatical at the Jet Propulsion Laboratory, Pasadena. He serves as a Permanent Lecturer for radar system engineering and for wave propagation For the Carl Cranz Series for Scientific Education. He is a Member of an Advisory Committee of the EU-Joint Research Centre (Ispra/Italy), and he is an advisor to the German Research Council (DFG), to the Federal German Ministry for Research and to industry in Germany. His research topics include radar, remote sensing, wave propagation and antennas. Dr. Wiesbeck has received a number of awards including the IEEE Millennium Medal. Since 2002, he has been a Member of the “Heidelberger Akademie der Wissenschaften.” He was a Member of the IEEE GRS-S AdCom from 1992–2000, Chairman of the GRS-S Awards Committee from 1994 to 1998, Executive Vice President IEEE GRS-S from 1998 to 1999, President IEEE GRS-S from 2000 to 2002, Associate Editor IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION from 1996 to 1999, past Treasurer of the IEEE German Section. He has been General Chairman of the 1988 Heinrich Hertz Centennial Symposium, the 1993 Conference on Microwaves and Optics (MIOP ’93) and he has been a member of scientific committees of many conferences.
